A 3–5 GHz FSK-UWB transmitter for Wireless Personal Healthcare applications

A 3–5 GHz FSK-UWB transmitter for Wireless Personal Healthcare applications

G Model AEUE-51285; No. of Pages 12 ARTICLE IN PRESS Int. J. Electron. Commun. (AEÜ) xxx (2014) xxx–xxx Contents lists available at ScienceDirect I...

1MB Sizes 0 Downloads 0 Views

G Model AEUE-51285; No. of Pages 12

ARTICLE IN PRESS Int. J. Electron. Commun. (AEÜ) xxx (2014) xxx–xxx

Contents lists available at ScienceDirect

International Journal of Electronics and Communications (AEÜ) journal homepage: www.elsevier.com/locate/aeue

A 3–5 GHz FSK-UWB transmitter for Wireless Personal Healthcare applications Hatem Trabelsi a,b,∗ , Imen Barraj b , Mohamed Masmoudi b a b

Department of Electrical Engineering, National Engineers School of Sfax, University of Sfax, B.P. 1173, Sfax 3038, Tunisia Laboratory of Micro-Electro Thermal Systems (METS), Tunisia

a r t i c l e

i n f o

Article history: Received 6 May 2014 Accepted 15 September 2014 Keywords: Dual band frequency-shift keying (FSK) Wideband resistive feedback power amplifier LC VCO Ultra-wideband (UWB) transmitter Wireless healthcare

a b s t r a c t A dual band frequency-shift keying (FSK) transmitter for low data rate, short range application, operating in the 3–5 GHz ultra-wideband is described in this paper. Chirped signal with FSK modulation enables direct conversion technique which simplifies the transceiver architecture. Two sub-band frequencies are used to encode the binary data employing chirped pulse generator. The front-end consists of an RF fast startup LCVCO, a ramp signal generator and a differential wideband resistive feedback power amplifier. All circuits are designed in 0.18 ␮m CMOS. The ramp generator produces linear rising ramp and can maintain a constant chirp range across PVT variation. The simulated chirp pulses show a Vpp of 0.26 V. The differential power amplifier achieves a P−1 dB of 4.4 dBm. It realizes a flat gain of 10.27 ± 0.03 dB across the dual band spectrum and a NF less than 2 dB. The transmitter maximum output power is in the −10.077 to −10.15 dBm range. The ramp generator, the pulse generator and the power amplifier consume currents of 0.95 mA, 1.6 mA and 7.5 mA respectively. The simulated PSD at the output of the transmitter meets FCC spectral mask requirement and the 500 MHz bandwidth constrain. The transmitter achieves energy efficiency of 0.6 nJ/bit at a data rate up to 3.2 Mbps. © 2014 Elsevier GmbH. All rights reserved.

1. Introduction Wireless Personal Healthcare technology is developing rapidly in recent years. With increasing number of aging population in most part of the world, healthcare system becomes a global issue [1]. The main objective of this wireless technology is to give an autonomous life, in their residence, to people suffering from various pathologies and handicaps. Medical remote monitoring systems are used to detect, even to prevent the occurrence of worrying or critical situations by the generation of messages and alarms on the situation of the person. Furthermore Wireless Personal Healthcare technology will improve healthcare for the rural and remote population by making specialty care more accessible.

∗ Corresponding author at: Department of Electrical Engineering, National Engineers School of Sfax, University of Sfax, B.P. 1173, Sfax 3038, Tunisia. Tel.: +216 74 274 088; fax: +216 74 677 657. E-mail addresses: [email protected] (H. Trabelsi), [email protected] (I. Barraj), [email protected] (M. Masmoudi).

In most hospitals, the number of medical staff is not sufficient. Doctors and nurses often work longer than they are expected, making them more prone to errors. With the advancement wireless technology, high performance wireless devices can be employed to improve the efficiency of hospital staff as well as to improve the comfort of patients. Fig. 1 is a block diagram of a typical sensor node. The sensing unit acquire physiological signals such as electroencephalography (EEG), electrocardiography (ECG), body temperature, blood pressure, etc., by individual sensors attached to the human body. The processing unit manages the communication protocols and processes the sensed signals. The data are transmitted with the transceiver unit through wireless link to a portable personal basestation or a remote healthcare center for analysis and diagnosis by medical professionals. Intelligence should be introduced at the network level to deal with network management and data interpretation. The main challenge for successful realization of the sensor nodes is the reduction of energy consumption to allow energy autonomy especially for implantable sensors. However, integration of the digital part and the RF part of the sensor node on the same chip using a

http://dx.doi.org/10.1016/j.aeue.2014.09.009 1434-8411/© 2014 Elsevier GmbH. All rights reserved.

Please cite this article in press as: Trabelsi H, et al. A 3–5 GHz FSK-UWB transmitter for Wireless Personal Healthcare applications. Int J Electron Commun (AEÜ) (2014), http://dx.doi.org/10.1016/j.aeue.2014.09.009

G Model

ARTICLE IN PRESS

AEUE-51285; No. of Pages 12 2

H. Trabelsi et al. / Int. J. Electron. Commun. (AEÜ) xxx (2014) xxx–xxx

Sensing unit Blood pressure Temperature ECG, EEG….

Power supply

Processing unit

Processor /storage

RF transceiver

Clock generation

Fig. 1. Block diagram of a sensor node on Human body.

cost efficient technology like CMOS and UWB standard are justified [2–3]. Current UWB systems are mainly based on two schemes: orthogonal frequency-division multiplexing (OFDM) or impulse radio (IR). OFDM-UWB may be able to provide a robust high data rate solution, but this comes at the expense of circuit complexity and power consumption due to the extensive digital signal processing required [4]. On the other hand, Impulse radio ultrawideband (IR-UWB) is well known for its low-duty cycle, high data-rate and high energy efficiency [5]. In this system, information is encoded in a very narrow pulse, which only occupies a small fraction of the symbol period. Besides the two mentioned schemes, a new dual band FSK modulation for the proposed UWB transceiver will be used. The use of dual band FSK modulation within the transmitted pulses eliminates the pulse width and amplitude constraints faced by conventional IR-UWB [6]. Low data rate IR-UWB communication system typically uses Gaussian pulses with extremely narrow width and high voltage swing to respect FCC spectrum mask requirement. The example below shows a peak-to-peak RF pulse amplitude calculation for a low data rate IR-UWB system. When considering a single band approach using a typical Gaussian pulse that occupies the 3–5 GHz band (band width B = 2 GHz), the half-amplitude pulse length is given by: tp ≈ (1.211/B) ≈ 0.6 ns. Considering a pulse repetition frequency (PRF) of 2 Mp/s, a peakto-peak RF pulse amplitude of more than 6.4 V is needed to reach the maximal allowed power density of −41.3 dBm/MHz. Thus, this high voltage requirement prevents such waveforms from being generated by low supply voltage CMOS technologies. However the proposed chirped spread spectrum FSK modulation technique with wider pulse width (75 ns) and smaller peak-to-peak RF pulse amplitude (260 mV) can achieve comparable PSD. Consequently, the reduced peak voltage, which is allowed by the proposed modulation scheme, has a direct influence on the power consumption of the radio-frequency front-end. The peak pulse voltage is furthermore increased when a passive band pass filter is used before the antenna in order to attenuate the unwanted emissions [7]. The goal of this work is to design a dual band FSK transmitter for low data rate, short range application, operating in the 3–5 GHz ultra-wideband (UWB) for Wireless Personal Healthcare application, which is easy to implement in silicon and is competitive in terms of power consumption. This paper is organized as follows: Section 2 presents the digital modulation and the working principles of the transceiver. The chirped waveform generator circuit and the wideband resistive feedback power amplifier circuit as well as simulation results of the transmitter are detailed in Section 3. Section 4 concludes this paper. 2. Transceiver architecture In this design we focus on modulation schemes that can be demodulated by simple non-coherent methods. This excludes

the use of phase-based modulations, which require sophisticated methods for signal detection and synchronization [8]. Different solutions for signaling scheme can be used. For on off keying (OOK) the BER degradation is usually acceptable for low-complexity receivers. However, OOK signaling raises some important issue for practical implementation. The strong dependence of the signal level with the data makes setting of a threshold value for optimum decision difficult [9]. Binary pulse position modulation (BPPM) has always been seen as one of the best candidates for IR-UWB. A benefit over OOK is the presence of a signal for any symbol. Moreover, the signal detection is much easier than for OOK since the energy in each half-bit periods is simply compared. Binary frequency shift keying (BFSK) non-coherent demodulation is equivalent to BPPM, since it relies on the detection of a signal in either of the two sub-bands instead of two half-bit periods for BPPM. BFSK features advantages over both OOK and PPM. First, for the same pulse rate, it is less affected by interpulse interference (IPI) and second, it only requires a zero threshold voltage for the decision operation [7]. For low data rate UWB wireless sensor application we need to use narrow width pulses with large peak amplitude to obtain the desired UWB spectrum. These pulses cannot be easily generated using advanced CMOS technologies which are not amenable to large voltage swings due to lower supply [6–10]. A chirp is a linear frequency modulated pulse. Chirped pulses or chirped spread spectrum (CSS) pulses is a spread spectrum system which can generate a sinusoidal signal whose frequency increases or decreases linearly over a certain amount of time. It could be thought of as sweeping the band at a very high speed. If the signal frequency rises from low to high it is called up-chirp signal, otherwise called down-chirp signal. The use of chirp modulation with wider pulse width and small pulse amplitude enables low complexity and low power transceiver implementation. The CSS solution can be used secured medical applications as it is very difficult to detect and intercept when operating at low power. This technique may also be used within wireless devices moving at high speeds, alarm systems for predicting vehicles’ collision, and vehicle-to-vehicle communication. Taking into account the previous analysis, we propose dual band BFSK as a modulation scheme. Instead of assigning fixed single tone carriers for BFSK the ‘1’s and ‘0’s are assigned fixed sub-bands of the lower UWB communications bandwidth (3.1–4.8 GHz). These subbands are 3.2–3.7 GHz for data bit ‘0’ and 4.0–4.5 GHz for data bit ‘1’ and are denoted as SUB1 and SUB2 respectively in the subsequent analysis. The transceiver architecture is shown in Fig. 2. The transmitting chain consists of a ramp generator that generates the control signal Vtune which in turn will control the dual band FSK chirped waveform generator. The FSK modulator generates dual band FSK modulated signal by switching between SUB1 and SUB2 signals depending on transmitter binary information input and Tx clock. The modulated signal is emitted by the antenna after it is amplified by a wide band power amplifier. The received dual band FSK pulses train is a random sequence of UWB impulses, each of which occupies either SUB1 or SUB2. The received signal is first amplified with a low noise amplifier (LNA) then down converted with the mixers using the quadrature I and Q sinusoidal tones produced by a quadrature voltage controlled oscillator (QVCO). A DAC is used to generate the control voltage for the QVCO. An off chip RF band pass filter between the T/R switch and antenna tend to worsen the system noise figure and increase the receiver complexity. However pass band filtering provided by both antenna and LNA can attenuate the out-of-band signals at the receiver input. Therefore RF band pass filter will not be used in this design. After the mixers a low pass sub-band-select filter, in each path, passes only the selected down-converted sub-band signal and suppresses the other sub-band. The filtered signals are amplified by

Please cite this article in press as: Trabelsi H, et al. A 3–5 GHz FSK-UWB transmitter for Wireless Personal Healthcare applications. Int J Electron Commun (AEÜ) (2014), http://dx.doi.org/10.1016/j.aeue.2014.09.009

G Model

ARTICLE IN PRESS

AEUE-51285; No. of Pages 12

H. Trabelsi et al. / Int. J. Electron. Commun. (AEÜ) xxx (2014) xxx–xxx

Low

Variable BW LP_Filters

3

Amplifiers

Mixers

Noise Antenna

Amplifier

Processor Dual band FSK demodulator

RX_output data

LO - I T/R Switch

QVCO

LO - Q QVCO control

DAC

Receiver Transmitter Power Amplifier

TX clock Dual band FSK Chirped waveform generator

Vtune Ramp generator

VCurrent cont Reset

Pulse generator control

Sub-band selection

2 bit TX_input data

Fig. 2. The overall FSK-UWB transceiver architecture.

a high gain amplifier. Finally a binary FSK non-coherent demodulator based on envelope detection will retrieve the digital transmitted data. 3. Transmitter design According to FCC masks restricts the UWB emission power must be under −41.3 dBm/MHz in the ranges of 3.1–10.6 GHz to reduce the potential interferences to existing applications and the side lobe suppression is required to be larger than 10 dB for the indoor system like wireless healthcare application [11]. The transmitter specifications are presented in Table 1. ADS (advanced design system) tool was used to simulate the transmitter. Fig. 3 shows the implementation of chirped signal

Table 1 Transmitter specifications. UWB operation freq range Channel BW Maximum data rate PRF Modulation scheme Sub-band Supply voltage Vpp (pulse voltage) Pulse width Transmitter output Technology Maximum range

Vout0

Txdata

PtRF_Pulse PORT1

DT

VtLFSR_DT SRC1

3–5 GHz 500 MHz 3.2 Mbps 3.2 MHz Dual band chirp FSK Chirp = 500 MHz SUB1: 3.2–3.7 GHz for ‘0’SUB2: 4–4.5 GHz for ‘1’ 1.8 V <0.3 V 75 ns −14.3 dBm CMOS 0.18 ␮m 15 m

VMult MULT1

R R1 Voutmod

VSum SUM1 Txdata

PtRF_Pulse PORT2 DT

VtLFSR_DT VMult SRC10 MULT2

R R6

Vout1 R R3

Fig. 3. Implementation of dual band FSK chirped signal generator.

Please cite this article in press as: Trabelsi H, et al. A 3–5 GHz FSK-UWB transmitter for Wireless Personal Healthcare applications. Int J Electron Commun (AEÜ) (2014), http://dx.doi.org/10.1016/j.aeue.2014.09.009

G Model

ARTICLE IN PRESS

AEUE-51285; No. of Pages 12 4

H. Trabelsi et al. / Int. J. Electron. Commun. (AEÜ) xxx (2014) xxx–xxx

of SUB1 signal, SUB2 signal and dual band FSK signal respectively. Fig. 5(f) shows FSK SUB1 and SUB2 in frequency domain for a pulse width tp of 75 ns and a PRF equal to 3.2 MHz. It is clear that the simulated power spectral density (PSD) meets FCC spectral mask requirement and the 500 MHz bandwidth constrain. For linear sweeping of the differential LC VCO frequency during chirped pulse generation, Vtune should have a linear relationship with time. We will consider in the following the ramp generator circuit.

Transmitter output frequency in GHz

4.5 SUB2

4 3.7 SUB1

3.2

3.1. Ramp generator circuit Fig. 6 illustrates the ramp generator circuit. The principle is to charge a capacitor with a constant current source. Capacitor C is the charging capacitor. Transistors M14 –M17 form a cascode current mirror and M18 is the current driver. M19 is used for stopping the current source when Tx clock is at logic 1. M20 is used to reset the capacitor. If the output impedance of the current source is neglected, Vtune is approximately given by (2):

t Transmit data

0

Tx clock

1 t

Vtune =

t

Ic (Cox W/2L) t= (VCurrent count − VT )2 t C C

with 0 ≤ t ≤

Chirped pulses

Tclock 2

(2)

where Tclock is the period of the Tx clock signal and VCurrent count is the voltage used to control the current Ic . For t = (Tclock /2), Vtune is directly proportional to Tx clock pulsewidth.

t

3.2. Chirped waveform generator circuit Fig. 4. Transmitter output frequency versus time with corresponding binary data and chirped pulses.

generator for SUB1 and SUB2. In Fig. 3 we use a PtRF pulse which is a pulse modulated RF source with frequency chirping. The chirped pulse is critical in this design. In order to study the mechanism of chirped pulses at system level we have used an ideal multiplier to multiply the chirped signal by Tx binary data and generate both sub-bands. We have determined the output signal in time and frequency domain. We will consider the non-linear effect in the design at transistor level of the transmitter and mainly when we simulate the IIP3 and the 1-dB compression point of the power amplifier. The carrier frequency at the start of the pulse is 3.2 GHz for SUB1 and 4 GHz for SUB2. The Chirp value represents the amount of frequency shift over the full, extended pulse width which is 500 MHz. Voutmod is the dual band FSK chirped signal. The generator will send SUB1 signal in case of data bit ‘0’ and SUB2 signal in case of data bit ‘1’. Fig. 4 shows transmitter pulse frequency versus time with SUB1 and SUB2 bandwidth limits, as well as binary data and corresponding chirped pulses signal. In this design one pulse per bit is transmitted. The −41.3 dB/MHz maximum transmitting power translates into a total carrier power (Ptx ) of −14.3 dBm, or 37.2 ␮W for a bandwidth (BW) of 500 MHz using the following Eq. (1): P(dBm/Mhz) = Ptx − 10 log BW

(1)

In 50  load this gives 0.12 V as a peak-to-peak pulse voltage amplitude. In order to compensate for losses in chip packaging a margin of 4 dB is added. This gives a transmitting power of approximately 100 ␮W which correspond to a peak-to-peak pulse voltage amplitude of 0.2 V in 50  load. Therefore minimum peak-to-peak pulse voltage amplitude of 0.2 V is required in order to meet sub-bands frequency limits and FCC mask requirement. For a maximum target pulse repetition rate of 3.2 Mp/s, the simulated chirp pulses in Fig. 5(a) shows a peak-to-peak pulse voltage amplitude of 0.2 V. Fig. 5(b, c and d), show the time representation

UWB pulse generator can be designed using different techniques. In [12] UWB pulses are generated by gated oscillators (e.g., tunnel diode or oscillator with output gate). Technique involving step-recovery diode used in conjunction with micro strip lines is proposed in [13]. These types of pulse generators are not easy for integration and not suitable for low-power, low-complexity and low-cost CMOS integrated technology. The approach in [14] based on a phaselock loop (PLL) provides a side lobe suppression of around 20 dB. However, the high DC current consumption and complexity of the circuit make it less attractive due to the large chip size. The conventional differential LCVCO is a good candidate for low to moderate data rate applications where low power consumption is the most critical design objective. In addition it can handle multi-band operation using capacitor bank and transmission gate switches. However this technique can achieve moderate spectral efficiency. By adding guard bands and increasing channel spacing, more out of band emissions may be tolerated for the proposed application. The fundamental idea is to create the SUB1 and SUB2 pulses, by selecting proper capacitors bank and by turning ON and OFF the oscillator using a controlled switch. The switch can satisfy power saving and oscillation switching. The use of switches helps minimizing the fall and rise time of the pulse. Resistors and switch resistances have been minimized to reduce the charging time for fast frequency hopping. The pulse generator in Fig. 7 is based on a cross-coupled transistor pair which is modeled by a non-linear negative conductance gm and an equivalent RLC parallel resonator. The output wave-form is approximately given by (3): VPG (t) = A(t) · cos(ω0 t − ϕ)

(3)

In Fig. 3 we use a PtRF pulse which is a pulse modulated RF source with frequency chirping. The pulse amplitude characteristics are defined using the Gaussian-shaped erf pulse function. Instead of the rise and fall portions being linear ramps, this source generates a pulse based on

Please cite this article in press as: Trabelsi H, et al. A 3–5 GHz FSK-UWB transmitter for Wireless Personal Healthcare applications. Int J Electron Commun (AEÜ) (2014), http://dx.doi.org/10.1016/j.aeue.2014.09.009

G Model

ARTICLE IN PRESS

AEUE-51285; No. of Pages 12

real(Vout0[0])

real(Txdata1[0])

H. Trabelsi et al. / Int. J. Electron. Commun. (AEÜ) xxx (2014) xxx–xxx

5

1.0 0.8 0.6 0.4 0.2 -0.0 -0.2 0.10

(a)

0.05

(b)

0.00 -0.05

real(Voutmod[0])

real(Vout1[0])

-0.10 0.10 0.05

(c)

0.00 -0.05 -0.10 0.10 0.05

(d)

0.00 -0.05 -0.10

0.0

0.3

0.6

0.9

1.2

1.5

time, usec

FSK SUB1 and SUB2 spectrum

Spectral density (dBm/Mhz)

-40

-50 -60 SUB 1

SUB 2

-70

-80

-90 -100

3.0

3.2

3.4

3.6

3.8

4.0 4.2 freq, GHz

4.4

4.6

4.8

5.0

Fig. 5. (a) Tx data, (b) Time representation of SUB1 signal, (c) Time representation of SUB2 signal, (d) Time representation of dual band FSK signal, (e) Chirp waveform, (f) FSK SUB1 and SUB2 in frequency domain for tp = 75 ns and PRF = 3.2 MHz.

the error function, giving a different shape to the rising and falling edges. By not having abrupt changes in slope, the pulse shape is more realistic and close to sinusoidal waveform. It is for this reason that the dual band FSK chirped signal generator in Fig. 3 can be implemented using the waveform generator in Fig. 7.

The fast startup LC voltage-controlled oscillator (VCO) in Fig. 7 is controlled by the ramp signal Vtune (shown in the previous section), Tx clock and TX data. The LC VCO core consists of a cross-coupled NMOS pair (M11 , M12 ) with W/L ratio of 84 ␮m/0.18 ␮m to provide a sufficient gm and an LC tank (L1 , L2 , C1 , Cva ). The technology’s typical

Please cite this article in press as: Trabelsi H, et al. A 3–5 GHz FSK-UWB transmitter for Wireless Personal Healthcare applications. Int J Electron Commun (AEÜ) (2014), http://dx.doi.org/10.1016/j.aeue.2014.09.009

G Model

ARTICLE IN PRESS

AEUE-51285; No. of Pages 12 6

H. Trabelsi et al. / Int. J. Electron. Commun. (AEÜ) xxx (2014) xxx–xxx Table 2 Ramp generator circuit components.

Vdd

M14

M15

M16

Vtune

Ic M18 C

Tx clock

Charging capacitor C

Resistor R18

W/L: 20.5 ␮m/0.18 ␮m

17 pF

100 

M17

R18

VCurrent cont

M14, M15, M16, M17, M18, M19 and M20

M20

Reset

M19

Fig. 6. Ramp generator circuit.

transconductance kn = ␮nCox equals approximately 40 ␮A/V2 for n-channel MOS transistors. a0 and a1 are 2-bits control for SUB1 and SUB2 center frequency selection. In the SUB1 and SUB2 selection circuit, three resistors are used in the switches controlled by 2-bits a0 and a1 . A resistor of 100 , connected to transistor gate, is used to reduce input current when the logic input level is high (switch is ON). Tow resistors of 800  connected to transistor source and drain are used to connect capacitors to GND when the switch is ON and to Vdd when the switch is OFF. Varactors enable an external analog voltage Vtune to cover the frequency range of 3.2–3.7 GHz (SUB1) for data bit ‘0’ and 4.0–4.5 GHz (SUB2) for data bit ‘1’. The oscillator start-up time has to be kept constant in order to maintain the same chirped frequency range (500 MHz) for SUB1 or SUB2. As the center frequency is different for the two subband, the tank resistance changes slightly resulting in different

oscillation amplitude and start-up time. To compensate for this, the trans-conductance of the cross-coupled transistors is controlled by Tx data signal through modification of the current source I1 resulting in similar start-up time for both sub-bands. The proposed chirped waveform generator dissipates dynamic current only during the pulse emission, for the remaining time of the Tx clock period, no dc current is supplied since I1 is switched OFF by SW1 and thus saving power. For SW1 width/length ratio of 1.5 ␮m/0.18 ␮m is used for both NMOS and PMOS transistors. The initial frequency of the chirp is given by (4): w0 =

1



(4)

LC0

where inductor L = L1 + L2 equals to 0.512 nH and C0 is the fixed capacitance that generates w0 . For example: w0 = 2 · 3.2 GHz for SUB1 ≥ C0 = 4.83 pF. For SUB2: C0 = 3.1 pF In order to observe a flat frequency response inside FCC mask limits, frequency should vary in a linear manner as a function of time. Eqs. (5) and (6) are used to determine how the total capacitance of the VCO must vary. wchirp = w0 + ˛t =



1 L(C0 + C )

L1

Vtune





C0 + C ≈ C0 1 −

Cvar

M09

Cvar

a0 M10

PGoutC2

C2

Cbank

M11

Sub-band selection Tx data VbiasSUB1





1

LC0 (1 − (2˛t/w0 )) (6)

2˛t w0



where 0 ≤ t ≤ tp

(7)

3.3. Ramp generator and chirped waveform generator simulation results

PGout+ a1



Eq. (7) shows that it is necessary to reduce the total capacity of the VCO with time to have a linear increase of the frequency.

C1

C1

˛t w0

where 0 ≤ t ≤ tp

L2

(5)

where C is the variable capacitance that gives rise to the chirp over the desired range. w0  ˛t ⇒ wchirp = w0 1 +

Vdd

where 0 ≤ t ≤ tp

M12 SW1 Tx clock

Tx clock I1 M08

VbiasSUB2 Fig. 7. Chirped waveform generator.

Fig. 8(a) shows time representation of ramp generator output signal (Vtune ) for different values of charging capacitor C. The value of the capacitor was increased from 10 pF to 20 pF. We notice that the curve is linear only for values greater than or equal to 17 pF. To obtain maximum amplitude at the output 17 pF is selected. The ramp generator operates properly because it achieves a ramp time of 75 ns and amplitude of 1.67 V which meet transmitter specifications. Table 2 shows the values of ramp generator components. Signals Vtune and Txclock in time domain for a charging capacitor of 17 pF are shown in Fig. 8(b). Table 3 shows the values of chirped waveform generator components. Spiral inductor L1 and L2 are designed with (W of 30 ␮m, nr of 1.5 and rad of 70.1 ␮m) which gives an inductor value of 0.256 nH. Fig. 9 shows simulation result of the Chirped waveform generator with the ramp generator. Fig. 9(a) and (b) shows spectrum for the frequency range of 3.2–3.7 GHz (SUB1) for data bit ‘0’ and 4.0–4.5 GHz (SUB2) for data bit ‘1’ respectively. The corresponding SUB1 pulse in time domain for tp = 75 ns also illustrated in Fig. 9(c).

Please cite this article in press as: Trabelsi H, et al. A 3–5 GHz FSK-UWB transmitter for Wireless Personal Healthcare applications. Int J Electron Commun (AEÜ) (2014), http://dx.doi.org/10.1016/j.aeue.2014.09.009

G Model

ARTICLE IN PRESS

AEUE-51285; No. of Pages 12

H. Trabelsi et al. / Int. J. Electron. Commun. (AEÜ) xxx (2014) xxx–xxx

Waveform of ramp generator circuit 2.0

time= 75.0nsec Vtune=1.67 C= 17.000000

Vtune, V

1.5

1.0

0.5

0.0 0

10

20

30

40

50

60

70

80

90

100

time, nsec

transistor causes the amplifier to show narrow band characteristics and the input matching degrades the NF rapidly with frequency. Another approach is to use a resistive shunt feedback network (Fig. 10(a)). Resistance RS is the source resistance, Rsf is the feedback resistor and RL is the load resistance. In this configuration the feedback network can be designed in such a way that it can provide the required match at both the input and output ends. Motivated by this advantage and good wideband behavior this architecture has been chosen and in the subsequent sections, after analyzing a simplified circuit, we propose a new circuit based on the shunt-series feedback topology. The resistive feedback is used to provide input and output impedance of R0 = 50 . To obtain this and from the small signal model of the common source stage (Fig. 10(a)) the required smallsignal transconductance gain of the NMOS transistor gm1 and Rsf can be obtained from the following expressions (8):

Ramp generator signals for a charging capacitor of 17 pF

Txclock, V

1.5 0.5 0.0

Cgs =

(8)

gm1 2ft

(9)

where ft is the gain-BW product. The bandwidth is obtained by (10):

1.5

RESET time

1.0

f3 dB =

0.5 0.0

Rsf = (1 − Av)R0

where Av is the open-loop voltage gain. The device input capacitance Cgs is proportional to its transconductance gm1 (9).

1.0

-0.5 2.0 Vtune, V

1 − Av ; R0

gm1 =

2.0

7

0

100

200

300

400

500

600

700

800

time, nsec Fig. 8. (a) Time representation of ramp generator output signal (Vtune ) for charging capacitor value swept over 10–20 pF, (b) time representation Vtune and Txclock for a charging capacitor of 17 pF.

The results are in accordance with FCC spectral mask requirement and the 500 MHz bandwidth constrain. By turning on the LCVCO only during the pulse generation, transmitter with high energy efficiency can be designed and constant power efficiency can be maintained across a wide range of data rate [6]. In order to maximize the link span between transmitter and receiver, the transmit signal is amplified by a power amplifier that drives the antenna through an external matching network [15]. 3.4. Wideband resistive feedback power amplifier Design of a broadband power amplifier with matched input impedance and flat gain, over the entire 3–5 Ghz band is a challenge. Wideband power amplifier can be implemented with several techniques. Distributed amplifiers [16] can produce large bandwidth. However, large chip area makes them unsuitable for wireless body medical telemetry application. Inductively degenerated common-source amplifier with a wideband input matching network can be used [17]. However, the inductor in the source of the Table 3 Chirped waveform generator components. M11 and M12

Varactor Cva

Spiral inductor L1 and L2

M09 and M10

W/L: 84 ␮m/0.18 ␮m

W: 45 ␮m

0.256 nH

W/L: 2 ␮m/0.18 ␮m

Current I1

C1

C2

SW1 NMOS and PMOS

1 mA (SUB1)1.6 mA (SUB2)

9.67 pF

6.2 pF

W/L: 1.5 ␮m/0.18 ␮m

gm1 2ft = Cgs (1 − Av ) (1 − Av )

(10)

This is twice the bandwidth of a simple common source stage (gm1(simple CS) = −(2Av/R0 ) and f3 dB(simple CS) = (ft /Av )) This configuration nearly doubled the bandwidth because the resistive feedback has provided an output impedance of R0 without losing significant output current in a physical loading resistance. To obtain the desired gain Av we need (1/2)gm1 so we get (1/2)Cgs and hence twice the bandwidth. The input and output impedance is given by (11): Zin =

Rsf + RL 1 + gm1 · RL

;

Zout =

Rsf + RS

(11)

1 + gm1 · RS

The voltage gain is given by (12): Av =

RL (1 − gm1 · Rsf ) RS + RL + Rsf + gm1 · RS · RL



RL (1 − gm1 · Rsf ) RL + Rsf

;

(Rsf  RS ) (12)

Shunt-series feedback with cascode common source topology can match the input impedance in a broad bandwidth and increase the gain [18]. It presents good input matching and low power consumption. The power amplifier is preceded by a differential LC voltagecontrolled oscillator. Therefore the PA has to be differential. Moreover, differential topology offers several advantages such as rejection of noise traveling in the substrate as well as the supply noise and attenuation of common mode signal [19]. The proposed wideband resistive feedback power amplifier is shown in Fig. 11. To provide the bandwidth requirement and the high power gain with low power consumption, both the resistive feedback and cascode topology have been used. Gain flatness can be obtain by resistive feedback topology, while usage of cascode topology can reduce some parasitic capacitance effects which degrade the gain performance at high frequency, and achieve higher gain for the same power consumption. The common-gate stage M2 provides isolation between the drain of M1 and output ports which eliminates the Miller effect from the common-source amplifier M1, thus allowing the design of larger gain-bandwidth product.

Please cite this article in press as: Trabelsi H, et al. A 3–5 GHz FSK-UWB transmitter for Wireless Personal Healthcare applications. Int J Electron Commun (AEÜ) (2014), http://dx.doi.org/10.1016/j.aeue.2014.09.009

G Model

ARTICLE IN PRESS

AEUE-51285; No. of Pages 12

H. Trabelsi et al. / Int. J. Electron. Commun. (AEÜ) xxx (2014) xxx–xxx

8

-35

-35

-40

-40

-50

-50

-60

-60

-70

-70

-80

-80 2.9

3.1

3.3

3.5

3.7

3.9

3.5

4.1

3.7

freq, GHz

(a)

3.9

4.1

4.5

4.7

4.9

freq, GHz

(b)

tp=75ns

4.3

0.15 0.10

PGout

0.05 0.00 -0.05 -0.10

(c)

-0.15 1.86

1.87

1.88

1.89

1.90

1.91

1.92

1.93

1.94

1.95

1.96

1.97

1.98

time, usec Fig. 9. Spectrum of the chirped waveform generator (a) for the frequency range of 3.2–3.7 GHz (SUB1 for data bit ‘0’), (b) for the frequency range of 4.0–4.5 GHz (SUB2 for data bit ‘1’), (c) corresponding SUB1 pulse in time domain.

For good efficiency and linearity the proposed circuit operates in Class-AB regime. The common source amplifier modified with the shunt feedback of resistor Rsf and capacitor Csf enhance bandwidth and improve wider input matching. Due to the presence of the resistive feedback the biasing point depend on the output voltage. So it cannot be set to the optimal point. Consequently, to achieve the desired gain, the circuit consumes more power than the case of having optimal biasing point. Csf is added to separate the output and the input DC voltage levels. Then, the optimal biasing point can be achieved. Therefore, the desirable gain is achieved with low power consumption. The inductor Lout is used for shunt peaking purpose at high frequency. The inductive load can provide a resonant peaking at the output when the amplifier starts to roll off at high frequencies and

equalize the power gain across the bandwidth by compensating the decreasing impedance of capacitance with the increase of frequency [11]. The series inductor Lg1 is used to further boost the gain at high frequencies and extend the bandwidth by resonating with the parasitic gate to source capacitance Cgs1 . The inductor Lg2 is chosen large enough to provide high impedance path to RF signal and to stop its flowing to bias circuit. Transistor M3 is used to bias the circuit. M3 and M1 form a current mirror source. Cb1 and Cb2 are input and output DC blocking capacitors. Fig. 12 shows the small signal equivalent circuit for the input part of the power amplifier. It contains Cb1 , Lg1 , Lg2 and Cgs1 . Components values are shown in Table 4. Cgs1 was calculated to be about 2.25 pF for ft of 11.25 GHz. This circuit is a linear phase filter. Its

Vdd RL RS

Vout RS Vs

Rsf

(a)

M1

Vs

Rsf Cgs

gm1vgs

RL

(b)

Fig. 10. (a) Amplifier with resistive shunt feedback for wideband matching, (b) small signal equivalent circuit.

Please cite this article in press as: Trabelsi H, et al. A 3–5 GHz FSK-UWB transmitter for Wireless Personal Healthcare applications. Int J Electron Commun (AEÜ) (2014), http://dx.doi.org/10.1016/j.aeue.2014.09.009

G Model

ARTICLE IN PRESS

AEUE-51285; No. of Pages 12

H. Trabelsi et al. / Int. J. Electron. Commun. (AEÜ) xxx (2014) xxx–xxx

9

Vdd Rb

Cb2

Cb2

M3

M6

Vout+ Vout-

M5

M2 Lg2 Vin+

Rb

Lout

Lout

Cb1 Lg1

Csf

Csf

Rsf

Rsf M1

PGout+=Vin+ PGout-=Vin-

Lg2 Cb1 Vin-

Lg1

M4 SW2 Tx clock

Tx clock

Fig. 11. Circuit of the proposed wideband resistive feedback power amplifier.

The simulated S-parameters are shown in Fig. 14. As can be seen, in-band power gain of 10.27 dB is achieved. However, across 500 MHz bandwidth the power gain is within 10.27 ± 0.03 dB, which is acceptable. The simulated in band S11 is less than −10.3 dB. The associated NF is shown in Fig. 15. The simulated in band noise figure NF is in the 1.95–2.06 dB range. Linearity is evaluated by ‘ADS HARMONIC BALANCE’ simulation. Fig. 16(a) shows that the power amplifier achieves a 1-dB compression point (P−1 dB ) of 4.4 dBm. Fig. 16(b) shows the input and output waveform at the compression point of 1 dB. This value of compression point ensures that the power amplifier is linear up to 4.4 dBm which is greater than the maximum transmitting power of −14.3 dBm. The third-order input intercept point IIP3 will be about 14.4 dBm. At PRF of 3.2 Mbps and a pulse width of 75 ns the power amplifier consumes an average power of 1.665 mW. Table 4 shows the values of the wideband resistive feedback power amplifier components. The ramp generator, the pulse generator and the power amplifier consume currents of 0.95 mA, 1.6 mA and 7.5 mA respectively. The transmitter consumes an average power of 1.93 mW at PRF

Cb1 Lg1 Vin Lg2

Cgs1

Fig. 12. Equivalent circuit at the input of the power amplifier.

response is desirable, due to its flat group delay. Such filter maximally preserves the wave shape of the input signal, since it delays all frequency components of the signal by the same amount. SW2 is used to enable/disable PA, in order to save power. It is controlled by the Tx clock signal from digital baseband. In order to compensate for losses in chip packaging and in antenna, a margin of 4 dB is added to the PA output power. The simulated in band (3.2–4.5 GHz) maximum output power is in the −10.077 to−10.15 dBm range (Fig. 13(a)). The simulated in band differential voltage gain is in the 8.47–10.32 dB range for both SUB1 and SUB2 signals (Fig. 13(b)).

m17

-10

m18

14 12

m14

10

dB(gain)

Output_power

-11 -12 -13

m12

8 6 4

m17 freq=3.200GHz Output_power=-10.077

-14

m18 freq=4.500GHz Output_power=-10.152

m12 freq=3.200GHz dB(gain)=8.472

2

-15

m14 freq=4.500GHz dB(gain)=10.322

0

1

2

3

4

5

6

7

1

2

3

freq, GHz

(a)

4

5

6

7

freq, GHz

(b)

Fig. 13. (a) Maximum output power in dBm of the power amplifier, (b) differential voltage gain of the power amplifier in the 1–7 GHz range.

Please cite this article in press as: Trabelsi H, et al. A 3–5 GHz FSK-UWB transmitter for Wireless Personal Healthcare applications. Int J Electron Commun (AEÜ) (2014), http://dx.doi.org/10.1016/j.aeue.2014.09.009

G Model

ARTICLE IN PRESS

AEUE-51285; No. of Pages 12 10

H. Trabelsi et al. / Int. J. Electron. Commun. (AEÜ) xxx (2014) xxx–xxx

m3 m2 freq=4.500GHz freq=3.200GHz dB(S(1,1))=-10.288 dB(S(1,1))=-15.403

m6

m7

m2

m6 m7 freq=3.200GHz freq=4.500GHz dB(S(2,1))=10.312 dB(S(2,1))=10.240

m3

m8

m13 m5 m4 freq=4.500GHz freq=3.200GHz dB(S(2,2))=-10.116 dB(S(2,2))=-9.849

m8 m13 freq=3.200GHz freq=4.500GHz dB(S(1,2))=-15.555 dB(S(1,2))=-15.282

m4

m5

Fig. 14. Power amplifier S-parameters.

of 3.2 Mbps with a pulse width of 75 ns. It requires 0.6 nJ/bit for transmitting. Table 5 shows a summary of simulated performance of the dual band FSK transmitter. The comparison with other reported state of the art designs is also presented. In [6] the ramp generator produces non-linear waveform and the charging capacitor needs to be initially charged to Vdd when Tx clock is low. The proposed ramp generator in this work produces linear rising ramp and consumes 0.95 mA only. However the charging capacitor does not need to be initially charged. The proposed ramp generator can maintain a constant chirp range across PVT variation using tow simple signals: VCurrent count to control the charging current and a reset signal to reset the capacitor. But in [6] a circuit containing a comparator and three switches is used to dynamically tune the discharge rate of the capacitor. This translates to more power consumption. The proposed LCVCO circuit is simpler compared to [6] and its output is an up-chirp signal with a pulse Vpp of 0.26 V. The LCVCO

consumes 1.6 mA while in [6] it is 17 mA. A wideband resistive feedback with cascode topology power amplifier is used to provide high power gain with low power consumption. It is linear up to 13.7 dBm and consumes 7.5 mA. In [6] the power is linear up to 2 dBm while consuming 22.3 mA. The proposed transmitter achieves energy efficiency of 0.6 nJ/bit while in [6] it is 0.77 nJ/bit. The use of a new dual band FSK modulation within the transmitted pulses eliminates the pulse width and amplitude constraints faced by conventional IR-UWB. By turning on the LCVCO and the power amplifier only during the pulse generation, transmitter with high energy efficiency can be designed. It is shown that the

Table 4 Power amplifier components. M1, M2, M4 and M5

M3 and M6

Inductor Lg1

W/L: 320 ␮m/0.18 ␮m W/L: 40 ␮m/0.18 ␮m 2.2 nH

Inductor Lg2

Inductor Lout

13 nH

4 nH

Cb1 and Cb2

Csf

Rsf

Rb

SW2 NMOS and PMOS

2 pF

1 pF

400 

1 k

W/L: 1.5 ␮m/0.18 ␮m

Fig. 15. Power amplifier noise figure.

Please cite this article in press as: Trabelsi H, et al. A 3–5 GHz FSK-UWB transmitter for Wireless Personal Healthcare applications. Int J Electron Commun (AEÜ) (2014), http://dx.doi.org/10.1016/j.aeue.2014.09.009

G Model

ARTICLE IN PRESS

AEUE-51285; No. of Pages 12

H. Trabelsi et al. / Int. J. Electron. Commun. (AEÜ) xxx (2014) xxx–xxx

11

(a) 2.0 1.5 1.0

ts(Vout), V ts(Vin), V

0.5 0.0 -0.5 -1.0 -1.5 -2.0 500

450

400

350

300

250

200

150

100

50

0

time, psec

(b)

Fig. 16. (a) 1-dB compression point of the power amplifier, (b) input and output waveform at the compression point.

proposed transmitter designed with dual band FSK modulator and wideband resistive feedback power amplifier achieves good performances over the 3–5 GHz bandwidth. The center frequency may have drifted due to PVT variation. If a coherent demodulator is used at the receiver, the shift in center frequency can cause bit error due to timing synchronization between transmitter and receiver. A non-coherent dual band FSK demodulator based on an envelope detection (or energy detection)

should be used in the receiver, where the amplitude information is more desired and the accuracy of pulse center frequency is less stringent. This technique relaxes center frequency tolerances, thus reducing bit error and allowing for reduced hardware complexity. The shift in center frequency could be tolerable in the proposed dual band FSK UWB system since it is much smaller than SUB1 or SUB2 band width.

Table 5 Summary of the transmitter performances and comparison with reported state of the art designs. Parameters

This work

[2]

[6]

[20]

[21]

[22]

RF freq Channel BW Modulation Typical data rate Linear up to output power Maximum Output power Supply voltage Technology Power consumption Energy efficiency

3–5 GHz 500 MHz Dual band FSK UWB 3.2 Mbps 13.7 dBm −10.15 dBm 1.8 V 0.18 ␮m CMOS 1.93 mW 0.6 nJ/bit

2.9–5.2 GHz 500 MHz FM UWB 0.1 Mbps – −10.2 dBm 1V 90 nm CMOS 0.9 mW 9 nJ/bit

3.15–3.9 GHz 400 MHz Dual band FSK UWB <20 Mbps 2 dBm −14 dBm 1.8 V 0.18 ␮m CMOS 14 mW 0.77 nJ/bit

3–5 GHz 500 MHz BPSK UWB 100 Mbps 8.5 dBm −14 dBm 1.2 V 0.13 ␮m CMOS 4.44 mW 44.4 pJ/bit

3–5 GHz 528 MHz OOK UWB 0.1 Mbps – −14.3 dBm 1.5 V 0.18 ␮m CMOS 0.0018 mW 18 pJ/bit

2.4 GHz – BFSK 0.125 Mbps – −5.2 dBm 1.8 V 0.18 ␮m CMOS 1.15 mW 9.2 nJ/bit

Please cite this article in press as: Trabelsi H, et al. A 3–5 GHz FSK-UWB transmitter for Wireless Personal Healthcare applications. Int J Electron Commun (AEÜ) (2014), http://dx.doi.org/10.1016/j.aeue.2014.09.009

G Model AEUE-51285; No. of Pages 12 12

ARTICLE IN PRESS H. Trabelsi et al. / Int. J. Electron. Commun. (AEÜ) xxx (2014) xxx–xxx

4. Conclusion In this paper, a new dual band FSK system architecture for an integrated, CMOS, 3–5 GHz impulse ultra-wideband transmitter suitable for medical telemetry sensor network applications is presented and modeled. In particular, the system is designed with a strong focus on reducing circuit complexity and power consumption. Lowering power is achieved through the use of dual band technique with a FSK modulation approach. Also by enabling the LCVCO and the power amplifier only during the pulse generation, constant power efficiency can be maintained. A fast startup LC voltage-controlled oscillator (VCO), a ramp signal generator and a low complexity differential wideband resistive feedback power amplifier are designed and simulations show good performance regarding transmitter specifications. Simulation results show advantages in overall performance compared to conventional IR-UWB transmitter topology. The transmitter is designed to communicate a maximum data rate of 3.2 Mbps with a pulse width of 75 ns. The results of this study validate dual band FSK transmitter as a low-complexity, low data rate UWB radio technology which enable low power, short range healthcare wireless sensor application. Implementation of a complete 3–5 GHz FSK-UWB transceiver with the processing and sensing unit using advanced CMOS technology is goals for the future research work. References [1] Stults BM. Preventive health care for the elderly. West J Med 1984;141:832–45. [2] Saputra N, Long JR. A fully-integrated, short-range, low data rate FMUWB transmitter in 90 nm CMOS. IEEE J Solid-State Circuits 2011;46(July (7)):1627–35. [3] Rabaey JM, Ammer J, Karalar T, Li S, Otis B, Sheets M, et al. PicoRadios for wireless sensor networks: the next challenge in ultra-low-power design. IEEE ISSCC Dig Tech Pap 2002:200–1. [4] Diao S, Zheng Y, Heng CH. A CMOS ultra low-power and highly efficient UWB-IR transmitter for WPAN applications. IEEE Trans Circuits Syst II 2009;56(March (3)):200–4. [5] Zheng Y, Tong Y, Ang CW, Xu Y-P, Yeoh WG, Lin F, et al. A CMOS carrier-less UWB transceiver for WPAN applications. IEEE ISSCC Dig Tech Pap 2006:378–87. [6] Nair MU. A low sir impulse-UWB transceiver utilizing chirp FSK in 0.18 ␮m CMOS. IEEE J Solid-State Circuits 2010;45(November (11)):2388–403. [7] Roovers R, Leenaerts DMW, Bergervoet J, Harish KS, van de Beek RCH, van der Weide G, et al. An interference-robust receiver for ultra-wideband radio in SiGe BiCMOS technology. IEEE J Solid-State Circuits 2005;40(December):2563–72. [8] Ghovanloo M, Najafi K. A wideband frequency-shift keying wireless link for inductively powered biomedical implants. IEEE Trans Circuits Syst 2004;51(December (12)):2374–83. [9] Gerrits JF, Farserotu JR, Long JR. Low-complexity ultrawide-band communications. IEEE Trans Circuits Syst II 2008;55(April (4)):329–33. [10] John FM, Michiel HL, Paul R, John RF, John RL. Principles and limitations of ultra-wideband FM communications systems. EURASIP J Appl Signal Process 2005:382–96. [11] FCC. Part 15.4: wireless medium access control (MAC) and physical layer (PHY) specifications for low-rate wireless personal area networks (WPANs); 2007. IEEE Std 802.15.4aTM -2007. [12] Zhao J, Maxey C, Narayanan A, Raman S. A SiGe BiCMOS ultra-wideband RFIC transmitter design for wireless sensor networks. In: Proc. IEEE Radio Wireless Conf. 2004. p. 215–8. [13] Han J, Nguyen C. A new ultra-wideband, ultra-short monocycle pulse generator with reduced ringing. IEEE Microw Wirel Compon Lett 2002;12(June):206–8. [14] Ryckaert J, Desset C, Fort A, Badaroglu M, De Heyn V, Wambacq P, et al. Ultrawideband transmitter for low-power wireless body area networks design and evaluation. IEEE Trans Circuits Syst I 2005;52(December (12)):2515–25.

[15] Mercier PP, Daly DC, Chandrakasan AP. An energy-efficient all-digital UWB transmitter employing dual capacitively-coupled pulse-shaping drivers. IEEE J Solid-State Circuits 2009;44(June (6)):1679–88. [16] Ballweber BM, Gupta R, Allstot DJ. A fully integrated 0.5–5.5-GHz CMOS distributed amplifier. IEEE J Solid-State Circuits 2000;35(February (2)):231–9. [17] Bevilacqua A, Niknejad AM. An ultra wide-band CMOS LNA for 3.1–10.6 GHz wireless receivers. ISSCC Dig Tech Pap 2004:382–533. [18] Kim C-W, Kang M-S, Anh PT, Kim H-T, Lee S-G. An ultra-wideband CMOS low-noise amplifier for 3–5 GHz UWB system. IEEE J Solid-State Circuits 2005;40(February (2)):544–7. [19] Ryckaert J, der Plas GV, Heyn VD, Desset C, Poucke BV, Craninckx J. A 0.65-to1.4 nJ/burst 3-to-10 GHz UWB all-digital TX in 90 nm CMOS for IEEE 802.15.4a. IEEE J Solid-State Circuits 2007;42(December (12)):2860–9. [20] Lingli X, Yumei H, Zhiliang H. A fully integrated BPSK amplitude and spectrum tunable transmitter for IR-UWB system. J Semicond 2009;30(January (1)), 015006-1–015006-5. [21] Tuan-Anh Phan. A 18-pJ/Pulse OOK CMOS transmitter for multiband UWB impulse radio. IEEE Microw Wirel Compon Lett 2007;17(September (9)):688–90. [22] Ayers J, Panitantum N, Mayaram K, Fiez TS. A 2.4 GHz wireless transceiver with 0.95 nJ/b link energy for multi-hop battery free wireless sensor networks. In: Symp. VLSI Circuits Dig. Tech. Papers. 2010. p. 29–30. Hatem Trabelsi was born in Sfax, Tunisia in 1969. He received the engineering degree in electrical engineering from the National School of Engineers of Tunis in 1993, the aggregation degree in 2001 and Ph.D. in electrical engineering from the National School of Engineers of Sfax in 2009. His doctoral research was mainly focused on design and implementation of ultra low power transceivers for Wireless Sensor Network. He is currently an Assistant Professor at the Electrical Engineering Department of the National School of Engineers of Sfax. He is a member of the research laboratory METS LAB ENIS, where he is working in the field of analog and RF circuit design for ultra-wideband systems. His research interests are singlechip CMOS transceivers, RF front-ends, Ultra wideband analog circuit design. Imen Barraj was born in Tunisia, in 1986. She received the Engineer Diploma in Electronics and Advanced Technologies from the Southern Private University of Sfax, Tunsia in 2010 and the Master degree in Electronic in 2011 from National Engineers School of Sfax, Sfax, Tunisia. She joints the Micro-Electro Thermals Systems (METS) research group since 2010. She is currently a Ph.D. student. Her current field in research is in radio architectures and design of Analog CMOS RF integrated circuits for Ultra WideBand transceivers.

Mohamed Masmoudi was born in Sfax, Tunisia, in 1961. He received the Engineer in electrical Engineering degree from the National Engineers School of Sfax, Sfax, Tunisia in 1985 and the PhD degree in Microelectronics from the Laboratory of Computer Sciences, Robotics and Microelectronics of Montpellier, Montpellier, France in 1989. From 1989 to 1994, he was an Associate Professor with the National Engineers School of Monastir, Monastir, Tunisia. Since 1995, he has been with the National Engineers School of Sfax, Sfax, Tunisia, where, since 1999, he has been a Professor engaged in developing Microelectronics in the engineering program of the university, and where he is also the Head of the Laboratory Electronics, Microtechnology and Communication. He is the author and coauthor of several papers in the Microelectronic field. He has been a reviewer for several journals. Dr. Masmoudi organised several international Conferences and has served on several technical program committees.

Please cite this article in press as: Trabelsi H, et al. A 3–5 GHz FSK-UWB transmitter for Wireless Personal Healthcare applications. Int J Electron Commun (AEÜ) (2014), http://dx.doi.org/10.1016/j.aeue.2014.09.009