A metasurface based low-profile reconfigurable antenna with pattern diversity

A metasurface based low-profile reconfigurable antenna with pattern diversity

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Journal Pre-proofs Regular paper A Metasurface Based Low-Profile Reconfigurable Antenna with Pattern Diversity Huy Hung Tran, Tuan Tu Le PII: DOI: Reference:

S1434-8411(19)32506-3 https://doi.org/10.1016/j.aeue.2019.153037 AEUE 153037

To appear in:

International Journal of Electronics and Communications

Received Date: Revised Date: Accepted Date:

2 October 2019 12 December 2019 13 December 2019

Please cite this article as: H.H. Tran, T.T. Le, A Metasurface Based Low-Profile Reconfigurable Antenna with Pattern Diversity, International Journal of Electronics and Communications (2019), doi: https://doi.org/10.1016/ j.aeue.2019.153037

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A Metasurface Based Low-Profile Reconfigurable Antenna with Pattern Diversity Huy Hung Tran1 and Tuan Tu Le2,3,* 1Division

of Electronics and Electrical Engineering, Dongguk University-Seoul, Seoul 100-715, South Korea

2Division

3Faculty

of Computational Physics, Institute for Computational Science, Ton Duc Thang University, Ho Chi Minh City, Vietnam

of Electrical and Electronics Engineering, Ton Duc Thang University, Ho Chi Minh City, Vietnam *Corresponding author: [email protected]

Abstract A low-profile reconfigurable antenna with a capability of generating multidirectional beams is presented in this paper. The design consists of a 4 × 4 unit cells metasurface (MS), a four-slotted ground plane, and a reconfigurable feeding network. By offsetting the slots from the center of MS, the unit cells of the MS can be excited with different phases and the main beam of the antenna can be steered to the desired direction. The pattern reconfigurablility is conveniently accomplished by electrically controlling the ON/OFF states of four p-i-n diodes on the feeding network to excite the proper slot. The final antenna has low profile of 0.04λo at the center operating frequency. The measurement indicates that the antenna has wide operating band of 10.4% by combining two adjacent resonances generated by the slot and the MS. Besides, the antenna exhibits four distinct beams with a peak gain value of 5.6 dBi. It is worth to note that compared to other pattern reconfigurable antennas in literature, the proposed design has advantages of low profile, wide operating bandwidth, and small number of p-i-n diodes. Keywords: metasurface (MS), pattern reconfigurable, wideband, slot

1. Introduction Recently, reconfigurable antenna has been extensively researched due to its flexible operation characteristics, which are suitable for various wireless communication platforms. In general, a reconfigurable antenna can be classified into four major categories including frequency [1], polarization [2], radiation pattern [3], and a combination of any of these properties [4, 5]. Among these types, pattern reconfigurable antennas are desirable for surveillance and tracking systems since their radiation pattern can be altered in several predefined directions. The first type of reconfigurablity is a dynamic switching the main beam shape, in which the radiation pattern is changed from boresight beam to conical beam [6–9] or monopolelike pattern [10, 11]. The second type is to change the radiation pattern in certain directions [12–21]. This paper focuses on reconfigurable antenna with multidirectional beam realization. For this purpose, the most common method is to employ reconfigurable parasitic element incorporated with primary radiating element [12–15]. However, using striplines [12], dipoles [13] or metallic walls [14] as parasitic element always requires for high antenna profile of greater than 0.2λo. In [15], the parasitic element based on four complementary split-ring resonators embedded in ground plane can help to achieve lower profile. Another method for multiple beam realizations is to excite the radiating element in different manners [16, 17]. Alternatively, using reconfigurable partially reflective surface (PRS) is also another effective way to produce multiple beams with high gain radiation [18–21]. However, the reconfigurable PRS has extremely large number of diodes and the antenna’s profile is always higher than 0.5λo. Recently, the use of metasurface (MS) structure has been demonstrated as a promising method to design beam-steering antennas [22–25]. In such designs, the MS is placed in close proximity with the radiating source, leading to the low antenna’s profile. In general, there are three ways to

achieve beam steering with the MS. The first method is to rotate the MS around the radiating patch [22]. The advantage of this technique is that the radiation pattern can be steered continuously, but it is hardly suitable in fast time-varying control systems due to the mechanical operation. The second method is to design reconfigurable MS as presented in [23]; however, it requires for large number of p-i-n diodes. To overcome these deficiencies, the third technique is proposed in [24, 25], in which the radiating source is shifted from the center of MS. To the best of authors’ knowledge, there is limited number of pattern reconfigurable antennas based on MS in open literature. In this paper, a MS based reconfigurable antenna with four different beam realizations is proposed. The antenna consist of 4 × 4 unit cell MS, four slots embedded in ground plane, and a feeding network. The beam steering principle is based on the excitation of each MS unit cell with different phase delays achieved by shifting the slots out of the center of MS. Then, with the aid of four p-i-n diodes, the pattern reconfigurability can be conveniently accomplished. The antenna is simulated and characterized using simulation tool high-frequency structure simulator (HFSS).

2. Antenna geometry Figure 1 illustrates the geometrical configuration of the proposed pattern reconfigurable antenna. The antenna consists of a MS layer and a feeding network, which are patterned on two FR-4 substrates with dielectric constant of 4.4 and loss tangent of 0.025. The MS is a lattice of 4 × 4 squared patches printed on top side of a 1.6-mm-thick substrate. A reconfigurable feeding network and a four-slotted ground plane are located on both sides of a 0.8-mm-thick substrate. To achieve good impedance matching for Mode-3 and Mode-4, the matching stubs are added to the feeding lines of Slot #3 and Slot #4.

The reconfigurable characteristic of the proposed antenna is attained by utilizing four p-i-n diodes of type MA4SPS402 with forward bias resistance of R1 = 5 Ω and reverse bias capacitance of C1 = 0.045 pF [26]. The diodes are designated as D1, D2, D3, and D4 and their polarity arrangements are as depicted in Figure 1. To provide dc control voltages, the diodes are biased with the aid of five 220 nH RF chokes and one 100 pF dc-blocking capacitor.

3. Antenna operation characteristics a. High gain and wideband operation characteristic It has been known that a single slot antenna will have narrow band and low gain operation. Better performances can be achieved by using MS structure since the slot coupled MS sustains two closely modes TM01 and anti-phase TM02. As the mechanisms have been thoroughly investigated in [27], we just briefly mentioned in this paper. According to this study, the principles for wideband and high gain can be described as follows: 

The gain enhancement is the consequence of larger effective aperture when using MS. The antenna with unloaded MS has only one radiating slot. While, the antenna with MS has multiple radiating slots formed by the gaps between the unit cells of MS.



The bandwidth enhancement can be achieved by generating two adjacent resonances. The first resonance is associated with the MS loaded slot antenna. The second resonance is from the surface wave propagating on the finite size MS. In other words, the second resonance is associated with the cavity effects formed by the MS and the ground plane.

The proposed antenna is designed to have operating frequency in the band from 5 to 6 GHz. The slot is opted for operation at around 5.2 GHz and its length (lslot) is chosen about halfwavelength at this frequency. Meanwhile, the resonance of the MS with internal excitation can be

considered as the resonance of a cavity (Lcavity) formed by the MS and the ground plane [30, 31]. According to these studies, the resonance of the surface wave propagating on the finite-sized MS can be quantitatively defined by the following equation: 𝛽𝑠𝑤 =

𝑚𝜋 𝐿𝑐𝑎𝑣𝑖𝑡𝑦

=

𝑚𝜋 ; 𝑁×𝑃

m = 1, 2, …

(1)

where βSW is the propagation constant, P is the periodicity and N represents the number of unit cells. Figure 2 shows the simulated dispersion diagram of a single MS unit cell at the first two eigenmodes, TM (transverse magnetic) and TE (transverse electric). The unit cell is setup with master and slave boundary conditions and the details for simulation setup can be found in [32]. The dispersion diagram shows how much phase shift the unit cell has at a given frequency. The data in Figure 2 shows that for example when the phase shift of the single unit cell is 90°, the intersection between the dispersion curves and the vertical lines representing βSWP = 90° indicates the surface wave resonant frequency of the MS, which is in the range from 5 to 6 GHz. The number of unit cell can be achieved by inserting βSWP = 90° into Equation (1). With m = 1, the number of unit cell is 2 × 2 and with m = 2, the number of unit cell is 4 × 4. Since the MS is the radiating aperture of the antenna, larger MS will have larger aperture or more radiating slots, resulting in higher gain radiation. In this design, we choose m = 2 to have high gain radiation. For better demonstration the wideband principle of the proposed design, Figure 3 presents the simulated antenna’s reflection coefficient for different values of slot’s length (lslot) and unit cell’s size (wo). It can be seen that tuning lslot has significant effect on the lower resonance, which increases as decreasing lslot. Meanwhile, the higher resonance is almost stable with the variation of lslot. The results confirm that the slot is critical for the lower band operation of the proposed antenna. Similar phenomenon can be observed for higher resonance when tuning wo, as illustrated in Figure 3(b). However, since the MS acts as an effective radiating aperture of the antenna, the

variation of wo also has significant impact on the lower operating frequency, which shifts upwards as decreasing wo. b. Beam switchable mechanism In order to steer the main beam of an antenna array to a particular angle (θ), the antenna elements are required to be excited with a progressive phase (∅) of: ∅ = 𝑘𝑑𝑠𝑖𝑛(𝜃) =

2𝜋𝑑𝑠𝑖𝑛(𝜃) 𝜆𝑔

(2)

where k is the wave number, λg is the guided wavelength, d is the unit cell dimension (d = P). For the MS antenna with center slot excitation, the adjacent unit cells will be excited with the similar phase. It means that the progressive phase condition will not be achieved and the resulting radiated beam is in broadside direction (θ = 0º). When the radiating slot is offset from the center, the corresponding MS cell in front of the slot will be excited first. Then, the other cells will be excited in sequence with different time delays. It causes the different phase delays among the unit cells of MS and thus, the main beam of the radiation pattern can be steered off the broadside direction. This mechanism has been thoroughly investigated in [25]. The radiation pattern reconfigurability of the proposed antenna can be attained by turning ON one diode while switching OFF the others, as illustrated in Table 1. Figure 4 shows the 3-D radiations of the antenna at 5.2 GHz in correspondence with each operating mode. It can be seen that the main beam is tilted off from the broadside to four different directions by exciting different slots. c. Key parameter studies Since the antenna is asymmetric along y-direction, the matching performance when operating in Mode-1 and Mode-2 will be different from Mode-3 and Mode-4. It is noted that due to the symmetry along x-direction, only results for Mode-1 and Mode-3 are chosen to present for brevity.

The optimization based on parameters sweep indicates that it’s difficult to achieve good reflection coefficient features for both Mode-1 and Mode-3. Here, the microstrip line is optimized to have the best performance for Mode-1. In that case, the matching of Mode-3 will become worse. To compensate it, additional stubs are introduced to the feed lines of Mode-3 and Mode-4, as depicted in Figure 1. By tuning the position (dstub) and the length (lstub) of these stubs, the matching performances of Mode-3 and Mode-4 are significantly improved. For demonstration, Figures 5 and 6 show the effect of dstub and lstub on the reflection coefficient of Mode-1 and Mode-3. It can be seen that these parameters have insignificant effect on the matching of Mode-1. In contrast, the matching of Mode-3 is considerably affected. By tuning dstub or lstub, the antenna input impedance can be adjusted conveniently and with proper values, good matching performance can be attained. The other critical in determining the operation of the proposed pattern reconfigurable antenna is the p-i-n diode type. It has been known that the diode with smaller bias capacitance when working in reverse state can provide better isolation than that with higher one. Therefore, the interference from unexcited feeding lines to the excited one will be minimized, resulting in better antenna performance. Besides, the diode with higher value of forward resistance will has more power loss, leading to lower gain radiation. For illustration, Figure 7 presents the antenna reflection coefficient of Mode-1 with different types of diode. The first case is the utilized diode in the proposed design, which is type MA4SPS402 with equivalent forward bias resistance and reverse bias capacitance of R1 = 5 Ω and C1 = 0.045 pF, respectively. In the second case, the diode with R2 = 5 Ω, C2 = 0.28 pF is employed. To note that this kind of diode might not be practical and the values of R2 and C2 are only used for better demonstration in this study. The last case utilizes the diode type MADP-042505-13060 with R3 = 0.83 Ω and C3 = 0.28 pF [33]. It observes a minor

effect of the forward bias resistance (R) on the |S11|. On the other hand, there is a strong impact of reverse bias capacitance (C) and better antenna performance can be obtained with smaller C.

4. Measurement results and discussion An antenna prototype was fabricated by using the printed circuit board (PCB) and the photographs are illustrated in Figure 8. The measurements were carried out with the aid of PNA Network Analyzer N5224A and Microwave Technologies Group [34]. There is a small difference between simulations and measurements, which might be the imperfections in measurement setup and fabricated tolerances. The simulated and measured matching performances of the proposed antenna are presented in Figure 9. It observes the common frequency range for all operating modes with reflection coefficient |S11| ≤ –10 dB of 10.4% (5.0–5.55 GHz). It is noted that the difference between the simulated and measured results is mainly come from the misalignment between the MS and the slots. Figure 10 provides the data for the antenna realized gain across the common band. The measured peak gain is about 5.6 dBi and the measured gain is always better than 3.8 dBi for all states. The antenna gain radiation patterns at 5.2 GHz are plotted in Figure 11. As observed, the main radiation beam of the antenna is steered in different directions for different modes. Based on the switching condition, the main beam is tilted at about ±30° off the broadside direction in the elevation planes of φ = 45° and 135°. In addition, the measured cross-polarization values are always 20-dB lower than co-polarization values. Finally, the measured performances of the proposed antenna at every operating mode are summarized and given in Table 2. Table 3 shows a performance comparison among reconfigurable antennas with multidirectional beam. It is noted that for the proposed design, the ground plane must be further

extended from the MS to have space for soldering the SMA connector, making the antenna’s overall size larger. In comparison with the dipole antennas [12, 13, 17] and the antennas with PRS [18, 20, 21], our design is significantly lower profile and wider operating BW (except for [17]). Using patch antennas in [14, 15] can achieve similar operating BW as our design, but higher antenna profile. Meanwhile, compared to the MS antennas [23, 24], our design has largest number of operating states with smallest number of p-i-n diodes. Besides, lower profile is also another advantage of the proposed antenna.

5. Conclusion The MS based pattern reconfigurable has been proposed and investigated. The main mechanism is by placing the slot out of the center of MS, the beam radiation can be directed to desired location. In the proposed design, four radiating slots are positioned at four different corners of the MS and thus, the antenna is able to radiate four distinctive beams. The reconfigurability can be attained with the aid of four p-i-n diodes. The measurements on fabricated antenna show a wide operating BW of 10.4% and a peak gain value of 5.6 dBi. With the aforementioned characteristics, the proposed antenna can be a good candidate for 5-GHZ WLAN applications.

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Table 1 Bias condition for different operating modes Operation

D1

D2

D3

D4

Mode-1 Mode-2 Mode-3 Mode-4

ON OFF OFF OFF

OFF ON OFF OFF

OFF OFF ON OFF

OFF OFF OFF ON

Beam direction (φ, θ) (135°, -30°) (45°, 30°) (135°, 30°) (45°, -30°)

Table 2 Measurement results of four operating modes Band BW Gain Beam direction Operation (GHz) (%) (dBi) (φ, θ) Mode-1 5.00–5.65 12.2 4.2–5.6 (135°, -30°) Mode-2 5.00–5.68 12.7 4.1–5.7 (45°, 30°) Mode-3 5.00–5.55 10.4 3.8–5.4 (135°, 30°) Mode-4 4.98–5.58 11.4 3.9–5.3 (45°, -30°)

Table 3 Comparison among pattern reconfigurable antennas with multidirectional beam Ref. [12] [13] [14] [15] [16] [17] [18] [20] [21] [23] [24]

Overall size (λo)

Antenna structure

No. of diodes

No. of states

Operating frequency (GHz) 2.45 2.40 3.60 2.45 2.45 1.50 5.60 5.50 5.50 2.45 0.90

BW (%)

Gain (dBi)

Efficiency (%)

8 4.1 10.8 2.4 10.5 58 4.3 3.6 1.8 <3 27

6.5 5.0 6.0 7.5 4.5 6.2 15.5 9.7 10.4 5.8 8.0

Not Given 45 80 Not Given 70 80 Not Given 75 Not Given Not Given Not Given

5.20

10.4

5.6

58

0.58 × 0.58 × 0.26 Dipole 20 5 1.22 × 1.22 × 0.28 Dipole 12 5 0.53 × 0.53 × 0.22 Patch 6 6 0.82 × 0.78 × 0.02 Patch 4 9 0.66 × 0.66 × 0.05 Patch 8 8 0.66 × 0.53 × 0.25 Dipole 30 3 2.81 × 2.81 × 0.68 Patch + PRS 72 9 1.83 × 1.83 × 0.58 Patch + PRS 144 9 2.75 × 2.75 × 0.50 Patch + PRS 72 10 0.65 × 0.65 × 0.10 Wire + MS 18 2 0.99 × 0.77 × 0.06 Slot + MS 6 3 0.97 × 0.97 × 0.04* Prop. Slot + MS 4 4 0.71 × 0.71 × 0.04** *: Overall size with extended ground plane for feasible fabrication **: Overall size without extended ground plane

y z

x

y Lcavity h2 Layer-1

Layer-2

h1

#3

Layer-3 #4

y

(+)

Mode-2

DC wires

Mode-1 wslot

Slot #2

#2

P

Side-view

x

#1

wo

lslot

lf

lw

Slot #1

dstub

D2 D1

y lstub D 3

x Slot #4

Slot #3 Mode-3

wp

Mode-4

(+)

Wg

(+)

RF chokes wstub

D4

ww wf capacitor

(-)

(+)

Figure 1 Geometry of the proposed pattern reconfigurable antenna. The optimized parameters: Wg = 55 mm, h1 = 0.8, h2 = 1.6, wo = 9.5, P = 10, lslot = 9.6, wslot = 1, lf = 18.2, wf = 1.4, dstub = 5, lstub = 4, wstub = 1, lw = 3, ww = 0.4, wp = 2 (unit: mm). 10

Frequency (GHz)

8 6 4 TM waves TE waves Light line

2 0

0

30

60

90

βP (degree)

Figure 2 Dispersion diagram of the MS unit cell.

120

150

180

0

|S11| (dB)

-10

-20

-30

-40 4.8

lslot = 9.2 mm lslot = 9.6 mm lslot = 10 mm 5.0

5.2

5.4

5.6

5.8

6.0

Frequency (GHz) (a) 0

|S11| (dB)

-10

-20

-30

-40 4.8

wo = 9.3 mm wo = 9.5 mm wo = 9.7 mm 5.0

5.2

5.4

5.6

5.8

6.0

Frequency (GHz) (b)

Figure 3 Simulated reflection coefficient of Mode-1 against the variations of (a) slot’s length lslot, and (b) cell’s size wo.

Mode-1

Mode-2

Mode-3

Mode-4

Figure 4 3-D radiation patterns of four different beams.

0

|S11| (dB)

-10

-20 dstub = 4 mm dstub = 5 mm dstub = 6 mm

-30

-40 4.8

5.0

5.2

5.4

5.6

5.8

6.0

Frequency (GHz) (a) 0

|S11| (dB)

-10

-20 dstub = 4 mm dstub = 5 mm dstub = 6 mm

-30

-40 4.8

5.0

5.2

5.4

5.6

5.8

6.0

Frequency (GHz) (b)

Figure 5 Simulated reflection coefficients of (a) Mode-1 and (b) Mode-3 against the variation of stub’s position, dstub.

0

|S11| (dB)

-10

-20 lstub = 3.0 mm lstub = 4.0 mm lstub = 5.0 mm

-30

-40 4.8

5.0

5.2

5.4

5.6

5.8

6.0

Frequency (GHz) (a) 0

|S11| (dB)

-10

-20 lstub = 3.0 mm lstub = 4.0 mm lstub = 5.0 mm

-30

-40 4.8

5.0

5.2

5.4

5.6

5.8

6.0

Frequency (GHz) (b)

Figure 6 Simulated reflection coefficients of (a) Mode-1 and (b) Mode-3 against the variation of stub’s length, lstub.

0

|S11| (dB)

-10

-20 R1 = 5Ω, C1 = 0.045 pF R2 = 5Ω, C2 = 0.28 pF R2 = 0.83Ω, C2 = 0.28 pF -30 4.8

5.0

5.2

5.4

5.6

5.8

6.0

Frequency (GHz)

Figure 7 Simulated reflection coefficients of Mode-1 for different p-i-n diode types.

Figure 8 Photographs of fabricated antenna.

0

|S11| (dB)

-10

-20

-30 4.8

Mode-1 (Sim.) Mode-1 (Mea.) Mode-2 (Sim.) Mode-2 (Mea.) 5.0

5.2

5.4

5.6

5.8

6.0

Frequency (GHz) (a) 0

|S11| (dB)

-10

-20

-30 4.8

Mode-3 (Sim.) Mode-3 (Mea.) Mode-4 (Sim.) Mode-4 (Mea.) 5.0

5.2

5.4

5.6

5.8

6.0

Frequency (GHz) (b)

Figure 9 Simulated and measured reflection coefficients of the proposed antenna for all operating modes: (a) Mode-1 and Mode-2, (b) Mode-3 and Mode-4.

Realized Peak Gain (dBi)

7.5

5.0

Mode-1 (Sim.) Mode-1 (Mea.) Mode-2 (Sim.) Mode-2 (Mea.)

2.5

0

4.8

5.0

5.2

5.4

5.6

5.8

6.0

5.8

6.0

Frequency (GHz) (a)

Realized Peak Gain (dBi)

7.5

5.0

Mode-3 (Sim.) Mode-3 (Mea.) Mode-4 (Sim.) Mode-4 (Mea.)

2.5

0

4.8

5.0

5.2

5.4

5.6

Frequency (GHz) (b)

Figure 10 Simulated and measured realized peak gain of the proposed antenna for all operating modes: (a) Mode-1 and Mode-2, (b) Mode-3 and Mode-4.

Normalized Gain (dB)

0

-5

-10

-15

-20 -150

Mode-1 (Sim.) Mode-1 (Mea.) Mode-3 (Sim.) Mode-3 (Mea.) -100

-50

0

50

100

150

50

100

150

Theta (deg) (a)

Normalized Gain (dB)

0

-5

-10 Mode-2 (Sim.) Mode-2 (Mea.) Mode-4 (Sim.) Mode-4 (Mea.)

-15

-20 -150

-100

-50

0

Theta (deg) (b)

Figure 11 Simulated and measured radiation patterns of the proposed antenna at 5.2 GHz for all operating modes.

Declaration of interests

☒ The authors declare that they have no known competing financial interests or personal relationships that could have appeared to influence the work reported in this paper.

☐The authors declare the following financial interests/personal relationships which may be considered as potential competing interests: