microwave delay line
J. A. KUSTERS, *E.G.H.LEAN,**J.L.PELLEGRIN,t AND H. J. SHAWtt
A microwave acoustic delay line ordinarily has three parts. These are electromagnetic coupling circuit, transducer, and delay medium. In the device considered here, these three parts are coalesced into a single, simple, unstructured block of material, which has been termed an integral delay line. This type of delay line is suitable for use in both waveguide and coax systems, and both cases are described.
Fig 1 shows schematically a block of high Q dielectric material located in a waveguide. It is well known that such a sample will have electromagnetic resonances, and that at the resonant frequencies power will be absorbed from the waveguide and will set up strong rf electromagnetic fieids in and around the sample. The resonant mode of most interest here is the TE, 6 mode which is generally the lowest frequency mode, and f or which the internal rf electric.field has the form of closed loops as indicated schematically in Fig 1. It has been shown that this mode can readily be critically coupled to the waveguide by adjusting the position of the shorting plunger shown in the figure. For high dielectric constant, the walls of the resonator present approximately open circuit boundary conditions, for which the tangential component of the rf electric field reaches its maximum values at the walls. If the dielectric material is also piezoelectric, the tangential rf field at the walls will produce mechanical forces on these walls. In Fig 1 the two opposing faces F, and F, are assumed to be polished flat to an accuracy of a small fraction of an acoustic wavelength, and these two faces are also assumed to be mutually parallel to the same order of accuracy. Under these circumstances the two faces can act as singlesurface acoustic transducers. The mechanical forces on these faces will excite acoustic waves which are essentially plane waves propagating inward. An rf electromagnetic pulse travelling down the waveguide, typically a microsecond or less in duration, will excite the electromagnetic resonator mode for this same length of time. During this interval, acoustic tone pulses having the same frequency as the input signal will be excited at faces F, and F,. At the end of this interval, the electromagnetic energy in the resonator will drop to zero, but we are left with the acoustic waves travelling inward from the faces. The wave packet from F, will proceed to face F, (and vice versa) with a transit time T = b/V, in which b is the spacing between F, and F, and
* Presently at Hewlett-Packard California, USA
* * Presently at IBM Corporation, Watson Research Center, Yorktown Heights, New York f Presently at Stanford Linear Accelerator University, Stanford, California
ti US Naval Reserves Training Corp, University of Utah, Salt Lake City, Utah 84112, USA 238
ULTRASONICS October 1969
Fig 1 Dielectric
delay line inserted
V is the acoustic wave velocity. These two wave packets will pass through each other in the central region of the crystal without interacting. On arriving at the opposite faces the acoustic waves re-excite the electromagnetic mode of the resonator which then radiates into the waveguide an electromagnetic signal pulse which is a time delayed replica of the original input signal. The dielectric resonator is able to perform the function of a delay medium because in the microwave bands the dimensions for electromagnetic resonance can also be compatible with the time delay requirements for some applications. There are various ways in which this mechanism can be exploited. We will first describe a basic experiment which demonstrates the principles and shows that the behaviour agrees well with the theory. Experiments to date have used singlecrystal lithium niobate material. This material has the four qualities which are necessary for efficient operation of integral delay lines, namely high dielectric constant, high dielectric Q, high piezoelectric coupling coefficient and high acoustic Q. Various crystalline orientations have been used. Fig 1 illustrates the conditions applying to one S-band experiment. The waveguide is a standard S-band rectangular guide. The sample is located in the centre of the waveguide cross-section. For a laboratory experiment it is completely sufficient tc simply place the resonator on top of a piece of polyfoam. As indicated in the figure, the Y axis of the crystal is parallel to the waveguide axis, the X axis of the crystal is parallel to the electric field of the waveguide propagating mode, and the two faces F, and F, are Y-cut crystal faces. In one experiment the dimensions a, b, c of Fig 1 were 375,346 and 541 mils, respectively. The resonant frequency was 3762MHz. Pulses were fed into the waveguide input on the left in Fig 1,
and a circulatdr located at the input was used to channel echoes to a microwave receiver in the standard fashion for single-port delay line systems. The adjustable short was positioned such that the resonator was critically coupled and an input pulse at the resonant frequency produced no electrically reflected signal at the receiver. The delayed echo chain at the receiver had the usual properties found with singleport delay lines in that all echoes are spaced by the same basic delay time as that which separates the first echo from the input pulse. This basic delay time for the present experiment was 2.5~s. This is the one-way transit time across the distance b of Fig 1. The transverse electric fields of the resonator mode at faces F, and F, lead to the excitation of shear waves polarized along the X crystal axis. The propagation velocity of such waves along the Y axis predicts the 2.5~s time delay observed experimentally. The total insertion loss for the first echo was 54dB, this being the ratio of waveguide input pulse power to first echo power at the receiver. This was composed of 16dB conversion loss and 22dB acoustic propagation loss. The conversion loss figure is the ratio of waveguide input power to acoustic power excited at faces F, and F, and is thus the one-way conversion loss at the two transducers presented by the faces. It must be realized that these two transducers are excited phase coherently and their contributions to the echo signal add numerically. The same conversion loss figure applies again to the reconversion of acoustic to electromagnetic energy at these two faces. Successive echoes differ from one another in amplitude on average by an amount approximately equal to the propagation loss quoted above. This is the attenuation suffered by a shear wave in traversing the length, b, of the crystal medium at room temperature. We see that without further change the device operates directly as a repetitive delay line with equally spaced pulse echoes. The unloaded Q of the resonator as viewed at the waveguide input terminals was 4750. This is the Q of the combined circuit consisting of the dielectric sample plus the short-circuited section of the waveguide behind the sample. For this configuration,then,the operating bandwidth of the delay line is that of a resonator having this Q. In analysing the conversion efficiency, the power dissipated in the resonatox is expressed in terms of the electromagnetic energy stored in the resonant mode and the unloaded Q of the resonator. The acoustic wave power generated can be expressed in terms of the resonator rf electric field at faces F, and F, and the piezoelectric coupling coefficient. The ratio of these two power values is the conversion efficiency. Note that the calculation of electromagnetic energy stored in the resonant mode must contain the energy contributions stored both inside and outside the crystal, which are typically comparable in magnitude. Calculated values of conversion efficiency are typically within 1dB of the measured values. Other crystalline orientations offer the possibility of increased conversion efficiency. If the crystal in Fig 1 is rotated about its X axis so that the Y axis makes an angle 6’ with the waveguide axis, the piezoelectric coupling coefficient changes. In carrying out this rotation we keep the sample face orientations unchanged, so that, for example, F, and F, continue to be normal to the waveguide axis, but now their normals make an angle 0 with the Y crystal axis. A computer calculation shows that the electromechanical coupling coefficient varies with 0 and, near .5 = 60”, it reaches a maximum value which should provide some 3x improvement in conversion efficiency. The integral delay line can be loop-coupled directly to a coax line, resulting in a system which is even simpler and
H f ield.
x Fig 2 Loop
coupled delay line
much smaller than the waveguide case. This is illustrated schematically in Fig 2. A simple non-resonant metal enclosure (not shown) surrounds the resonator for rf shading. Critical coupling to the coax line is achieved by varying the spacing between the loop and face of the resonator. For the sample of the Fig 1, the axis of the coupling loop is parallel to the 2 crystal axis. When operated in this way, the same sample as discussed above gave essentially the same delay characteristics as when used in the waveguide. Total insertion loss was 1dB lower than with the waveguide coupling. The dielectric resonator has external rf fields, and in Fig 2 the external rf magnetic field for the resonator of Fig 1 is indicated by dashed lines. It has a dipolar shape. A series of resonators placed so that adjacent Z-cut faces are in close proximity are mutually coupled by their external fields. If such an array is used to replace the single resonator of Fig 1, the tuning bandwidth of the delay line is found experimentally to be increased. By tuning bandwidth we refer to the ability to tune the delay line over a range of frequencies by moving the shorting plunger. Using four coupled resonators a tuning bandwidth of 2% at 38OOMHzwas observed. Increases in instantaneous bandwidth also are possible using coupled resonators. Systems of coupled resonators in which an end resonator is fed by a coupling loop behave like classical coupled resonators, and are being studied as an approach to delay lines of increased instantaneous bandwidth. In summary, the integral delay line is a simple block of crystalline material which, if placed near a coupling loop in a coax system, or placed within a waveguide system, produces delayed echoes with reasonable conversion efficiency. It is very rugged, small and requires no critical adjustment. It may also be free of high field breakdown effects experienced with microwave delay lines using thin film transducers.
ACKNOWLEDGEMENTS The authors wish to acknowledge helpful consultations with Dr D. K. Winslow. The work was supported by the US Air Force of Aerospace Research and was carried out at the Microwave Laboratory, W. W. Hansen Laboratories of Physics, Stanford University, Stanford, California.
ULTRASONICS October 1969