Infrared Phys. Technol. Vol. 35, No. 213, pp. 463-416, 1994 Copyright 0 1994 Ekvier Science Ltd Printed in Great Britain. All rights reserved 1350~4495/94 $6.00+ 0.00
ANTENNA-COUPLED SUPERCONDUCTING CONTACTS SUBMILLIMETER AND FIR ASTRONOMY
H. ROTHBRMEL,’ B. PLATHNER,’ K. H. GUNDLACH,’ M. AOYAGI) and S. TAKADA~ ‘MPI f. extraterrestrische Physik, 8046 Garching, Germany, ‘IRAM, 38406 Saint Martin D’Heres, France and ‘Electrotech. Lab. l-l-4 Umezono, Tsukuba-shi Ibaraki, 305 Japan (Received
25 June 1993)
Abstract-Advantages and restrictions of an antenna coupled superconductor-isolator-superconductor (SIS) detector are discussed at the example of a 345 GHz open structure receiver. At submillimeter wavelengths an antenna coupled mixer is cost effective and provides ultimate bandwidth. Heterodyne systems with large IF bandwidth can challenge )He bolometers for continuum work. The status of open structure mixers in astronomy is presented. A noise budget based on a mathematical receiver model is in agreement with the measured receiver noise. An antenna coupled SIS detector in a dipole-lens combination has less beam efficiency than a waveguide mounting. As its sensitivity is equally good or better it makes up for its lack in efficiency by a higher transmission. If the junction capacitance is resonated out by a superconducting matching circuit, SIS mixers can approach the quantum noise limit. Fully assembled SIS receivers work at a few times this limit due to unavoidable losses in the signal path. Their application at frequencies up to 1 THz would be appealing for astronomy. A key role in this effort play efficient superconducting matching circuits which can handle the large static capacitance of junctions at operating frequencies much higher than the RC limited corner frequency. Niobuimnitride is a superconductor which exceeds the frequency limit of Niobium. The prospects of this material are addressed and first results are presented.
“Antennas, our electronic eyes and ears”“) is a nice motto to start a talk on antenna coupled detectors. Since the invention of the antenna (Heinrich Hertz 1886) a great variety of antennas has envolved. Kraus(‘) points out that radiation occurs by acceleration of electrical charge. Examples are: an oscillating charge, a charge moving along a curve and charge reflected at the end of a wire. The radiation is directed perpendicular to the acceleration. Receiving antennas are understood by the principle of reciprocity. In this context the antenna forms the interface between the focused radiation from an astronomical telescope and a detector which can be a bolometer, a photoconductor, a photovoltaic cell, a rectifier or a nonlinear superconducting device. As decisive difference to a detector mounted on an absorber, antenna-coupling restricts detection to one polarisation and one spatial mode. Antenna-coupling is governed by the antenna theorem: Aef.R=l’
where Aerr is the effective area, R is equivalent width of the main lobe in steradians and 1 is the wavelength. The antenna theorem quantifies the well known fact that an antenna with large surface, like a radio telescope, has a very narrow pattern, and the corresponding angular resolution increases for a given antenna linearly with frequency. II.
It is out of discussion that heterodyne detection is far superior to other methods if highest spectral and spatial resolution is required. For continuum sources (QSOs, dust etc.) ‘He cooled bolometers have lower detection limits because they measure in a bandwidth of 30-60 GHz and pick up both polarizations. In principle they also can couple to a number of spatial modes if a resolved source is pointed but this observing mode, providing even better photon statistics, is not 463
H. ROTHERMEL et al Table 1. Calculated detection limit for continuum radiation from point sources for a 345 GHz SIS receiver [this work] and a ‘He bolometer”) both used at the 30 m telescope on Pica Veleta. The limit is given in mJy for S/N = 1 and 1s of integration time. Single polarisation measurement, chopping, 100 K DSB receiver noise, Zenit transmission r = 0.3 and effective collecting areas of 70 and 35 m* at 45” and 90” antenna elevation are assumed for the heterodyne measurement
NEP [WI Bandwidth GHz
0.1 1 2 10 Bolometer
1.1 3.6 5.1 1.1
X x x X
10-16 lo-l6 IO_‘6 10-15
9.2 2.9 4.1 9.2
x x x x
IO-” 10mt6 10-16 lo-l6
1630 515 364 163
2630 831 587 263
> 20 GHz
standard when using a large radiotelescope. Table 1 shows the calculated detection limit of a 345 GHz heterodyne receiver in comparison to 3He bolometer working at the same wavelength. The heterodyne system breaks even at 2 GHz IF-bandwidth for 1 polarization and at 1.4 GHz bandwidth for dual polarization. Right now, with the heterodyne receiver, the continuum signal is detected with a single quadratic detector connected to an unequalized signal of 800 MHz bandwidth. Since the detector responds only to the highest peak in the spectrum the effective bandwidth of our continuum measurement is roughly 100 MHz. Obviously a careful equalization of the IF-spectrum already would pay in a large number of additional pointing sources. The design of a cryogenic IF-amplifier with 10 GHz bandwidth would lead to an attractive continuum receiver. Figure 1 shows a synoptic bandwidth of an open structure SIS mixer >24 GHz, much larger than the above IF-bandwidth. With the bandwidth-gain product of cooled HEMTs approaching 300 GHz,@) sufficient gain is available to design a low noise amplifier of 10 GHz bandwidth. In the atmospheric window around 260 GHz the detection limit of the bolometer is 40 mJy. The system works there in a synoptic bandwidth of 60 GHz and probably is unbeatable in sensitivity. It should be mentioned that antenna-coupling is not restricted to heterodyne systems. Bi bolometers(4.5) are used routinely for measurement of antenna patterns in the mm wavelength range and it is quite possible that 1 He cooled bolometers will use antenna coupling in astronomy in cases where the loss of 1 polarisation is desirable.
Fig. I. Normalized IF-signal from a 290 K load at the receiver entrance versus backplane position. The upper curve was measured at 333 GHz, the curve in the middle at 345 GHz, the lower curve at 357 GHz. With a backplane position of 100 nm the mixer gain is at its maximum value for all three frequencies, indicating an untuned frequency range of z 24 GHz.
Antenna-coupled Table 2. Double LO
In the millimeter
side band receiver noise temperatures measured in 50 MHz bandwidth installation at the 30m telescope on Pica Veleta, Spain 324 134
during 360 149
are made in waveguide
antenna provides a low sidelobe level and good main-beam efficiency. In the submillimeter range machining of waveguides and the corresponding substrates is cumbersome and surface losses and reflections at discontinuities become significant. Open structure designs are attractive for wavelengths below one mm because photolithography can produce small structures with high precision. These systems are convenient and cost effective because the mixer can be made as a fully integrated system consisting of antenna, matching circuit, SIS junction and IF-diplexer. Only the IF-line has to be connected. Blundell and Tong @)list as much as 8 open structure projects for frequencies between 115 and 761 GHz. There is no room to review all of them. It is problematic to compare laboratory results because an experimental receiver may work only at selected frequencies, its sidelobe level may be high or the receiver noise temperatures given may omit the transmission and emissivity of vital components in the signal path such as the LO- injection or beam forming lens. Beam matching, integrity to acoustical noise or electromagnetic interference, gain stability over many hours of integration time and a reliable, easily tunable local oscillator and phaselock system count as much as record setting numbers for noise temperature, if it comes to a serious effort at a radiotelescope. For an overview about the status of SIS receivers at submillimeter wavelengths we present only a few systems which are facility instruments or have been used for astronomy. The noise temperatures in Table 2 are a subset of more than 50 frequency tunings during an observing period at the 30m IRAM telescope in Jan. 93.“’ For the operation at the telescope a Macor collimator had been added on the 4 K plate in front of the open structure mixer in order to suppress sidelobes and scattered radiation. In addition a plastic lens of 100 mm focal length was needed to match the beam waists of telescope and mixer. in the laboratory (Fig. 11). This is why the values in table 2 are - 10 K higher than calibrations Nevertheless the collimation pays by a better overall efficiency taking full advantage of the telescope area and directing most of the receiver pattern to the cold sky. Figure 2 shows a comparison to corresponding systems installed at telescopes on Mauna Kea (Hawaii).
Fig. 2. A comparison of DSB receiver noise temperature of SIS receivers installed at or having been installed at 3 major submillimeter telescopes. (*) This work, (B) open structure receiver,(‘O) (W) waveguide receiver,““) (U) waveguide receivers.i3’)
H. ROTHERMEL Ed al.
345200 Rest Fig. 3. Deep integration
at a carbon star (IRC + 10216) using a 350 GHz open structure the IRAM 30 m Antenna on Pica Veleta, Spain.(*r
SIS receiver at
Astronomical sources observed Figure 3 shows a deep integration
before by other systems are a touchstone for performance. at a standard source. (*)At 4 times higher frequency resolution the signal to noise ratio is 2 times better than what can be found in the literature”’ and the base line is flat as shown without manipulation. Figure 4 shows a quasar measured to our knowledge the first time at a wavelength shorter than 1 mm. The 30 m antenna is the largest for submillimeter wavelengths but a 15 and a 10 m telescope in Hawaii have comparable collecting areas due to a higher surface accuracy. Hence the sensitivity on the sky is due to receiver sensitivity and stability. The large antenna pays in an angular resolution of 9”, considerably higher than what can be done elsewhere (14” and 21”). Angular resolution is beneficial e.g. in study of outflow sources and for mapping of external galaxies. 0528+134 Chan.
1 Rx345G 2
0.2 0.1 0 -0.1
Fig. 4. A Quasar (PKS 0528) with reshift z = 2.07, detected Quasars are important for pointing and, providing molecular found, for cosmological studies. The radiation from the Quasar, channel, is clearly above the background in each of
probably for the first time at 345 GHz.(‘) lines or atomic fine-structure lines can be added in the bandwidth of the continuum five 10 s on-source integrations.
superconducting contacts TECHNOLOGY
A radioastronomical receiver in open structure technique has the following major components: Local oscillator (LO), LO injection, cryogenic system, beam forming optics, feed antenna for the mixer element, matching circuit, the mixer element itself, diplexing filters for the IF- and bias circuitry, the cryogenic IF-amplifier and finally the magnetic coil for suppression of the Josephson effect. We address only what is of interest in context with antenna coupling and low noise. 1. LO injection and Dewar windows The beamsplitter for LO-injection and the Dewar window (Fig. 12) are at ambient temperature. Their insertion loss enters the noise budget twice: as a loss in signal and as source of additional noise through thermal emission. As the receiver noise-temperature is far below 300 K (Table 2), transmission is crucial. Only 2% of the LO is coupled in by the beamsplitter, corresponding to 98% transmission in the signal path. The Dewar window, made from low loss Polyethylene, measures 1.5 wavelengths. A Fabry-Perot resonance removes its surface reflection. 2. Beam forming optics and beam eficiency The pattern of the fully assembled mixer has to fit unvignetted through the Dewar window. Ideally it should match the f/l0 beam of a telescope without additional optics. So far it was not possible to create a narrow enough pattern with an antenna array directly. All successful open structure SIS receivers rely on one or the other kind of substrate lens (Fig. 5) to form the pattem.(“-“) The antenna-lens combination determines the beam efficiency, a crucial number which governs essentially the receiver noise (Fig. 12). In the case of a transmitter the beam efficiency is the ratio of power radiated through the mainlobe of the antenna over total power, radiated into 4n solid angle. In the case of a receiver, from the total incident power only the fraction, given by the beam efficiency, is available to the mixer due to reciprocity. In order to calculate the beam efficiency for our mixer (Fig. 5), we would have to know the angular pattern of an antenna deposited on the surface of a medium. The beam efficiency would be this pattern integrated inside the acceptance cone of the lens divided by the total integrated pattern. Fortunately the contour of the lens is positioned in the far-field for our geometry (Fig. 5), so that radiation vignetted by the aperture of the lens and surface reflections do not feed back on the antenna and can be treated separately. Two authors(‘6T17) present a calculated pattern of a short dipole on a dielectric surface.
Fig. 5. Cross section of an open structure SIS-mixer. (“.‘*) The substrate with antenna and junction deposited on it is attached to the rear of an substrate lens. The lens, made of crystal quartz, has an elliptic contour.
H. ROTHERMEL et al.
Fig. 6. Normalized bias current injected by the 345 GHz local oscillator for 4.2 mV bias voltage at a 2 junction array as a function of the backplane position. Dashed lines apply if the partition of forward/ backward power is proportional to n 3,(w’) dashed dotted lines if partition is proportional to the refractive index n.
The result is of great interest for the partition of powers radiated into the medium and the empty space behind. On the axis perpendicular to the dielectric surface the prediction is:
where S, and S, are power densities inside and outside of the medium, t, is the relative dielectric constant and n the refractive index. For crystal quartz (n = 2.105) one expects that 9% of the radiated power leaks into the half space behind a substrate lens. A measurement (Fig. 6) indicates that this is unrealistic. Moving the conducting backplane (Fig. 6) backward, a series of maxima and minima can be observed due to constructive and destructive interference of power radiated into the empty space behind the substrate. As d.c.-bias current is proportional to LO-power for moderate amplitude, no more than +9% deviation is expected according to the above relation. The actual deviations are f35%. A large modulation could be understood if the backplane position contributes to the match between antenna and SIS junction, but this is not convincing because (i) the junction is matched by a very low impedance transmission line on the substrate, (ii) the fringe pattern is very consistent all over a tuning range of 24 GHz and (iii) is observed with the same amplitude for a variety of antenna styles, junction areas and lens styles. The measured fringe pattern can be made plausible looking at the power POcarried by a wave as a function of the amplitude U,, in free space:
where 2, = 377 R is the impedance of free space and x is the distance to the surface, and the same quantities in a medium:
where n is the refractive
along the axis perpendicular
the amplitudes uo (0) = u, (0)
on the surface equal:
Fig. 7. Layout of a planar dipol antenna in between bandstop filters.
one gets an estimate for the partition of power radiated into the medium and into free space for all distances: P, = Pan
and for the total power P = P, + PO= (n + l)P,
In the case of crystal quartz P,, = P/(2.105 + 1) = 0.32 is in reasonable agreement with measurement (Fig. 6). 3. Planar antenna A variety of planar antennas are used in combination with substrate lenses: Planar spirals,(“) double dipole slots,Q8)double dipoles(“) and log. periodic patterns.(4’ We can show that a simple broadband dipole is sufficient for a reasonably good illumination of the aperture of the substrate lens. There may be better solutions but they need careful evaluation on the level of scaled models and good control of process parameters such as junction area, critical current density, penetration depth in superconducting films etc. Figure 7 shows an example for the layout of a planar antenna. This particular layout is not appealing, because it was found empirically, but in comparison to alternative designs according to handbooks, this works best. The design history is as follows: a PbBi-InO-PbBi SIS junction without matching circuit, designed for a 260 GHz waveguide mixer,u9’ was used first. It already provided a DSB receiver noise temperature of 200 K in the 350 GHz open structure mixer.(13’Next the structure was analyzed as a scaled model (*O) for its antenna impedance. Finally a superconducting matching circuit was designed”” and the antenna was repeated in Nb-Al-Alox-Nb technology leading to temperatures < 100 K. The antenna works roughly like follows: the inner two patches form a dipole, the outer periodic structures a bandstop for the input frequency, diplexing the IF-signal and the bias voltage. To understand the filter, one has to keep in mind that a metallic substrate holder, in a short distance behind the substrate, makes the counter electrode for a suspended stripline geometry. The filter cannot interfere with the angular pattern of the planar dipole because (i) the bandstop filter carries almost no current at the input frequency and (ii) the field energy is confined to a narrow gap in between the substrate holder and the metalization on the substrate. 4. Matching circuit
Nb-Al-Alox-Nb junctions with good I/V curves can be made for critical current densities up to 10 kA/cmZ. The insulating barrier measures only 10 atomic layers, leading to a large specific capacitance (Table 3). The RC product given by the junction capacitance and the junction normal resistance defines a corner frequency of N 100 GHz which does not depend on the junction area Table 3. Material constants Niobium film
(T(4.2 K) spec. jet. cap.
9K 2.84 mV 85 nm 1.57 x lO’(fh-’ 60 fF/pm*
Nb-Nitride film 14K 5mV 300 nm 3 x 106(Rm)120 lF/pm*
H. ROTHERMEL et al.
\ \ , -3
Fig. 8. Angular pattern of the receiver in telescope configuration including a beam matching polyethylene lens, measured with a chopped cold load at 345 GHz LO frequency. Solid line: E-plane cut, dashed dotted line: H-plane cut, vertical lines: edge of the secondary mirror.
t I Nb wiring layer II Si O2 insulatton III Nb botton layer fused
14OOnm+--j 100 nm 160 nm 120 nm
Fig. 9. Matching circuit (z‘) for a series array of 2 N&Al-Alox-Nb junctions of 2 pm* area each. The junctions are defined at a distance >1/2 away from the center on patches of the planar dipole. (a) Magnified top view of the antenna with the junction array in stripline connection. (b) Magnified detail showing the junction in top view. (c) Magnified cross section of the junction covered by a wiring layer (I) forming with the SiO, layer (II) and the bottom layer (III) a stripline configuration.
Antenna-coupled superconducting contacts because,
with ,larger area, the junction
operation above the corner frequency a resonant removes the capacitance and matches the junction
rises at the same rate as the capacitance. matching circuit becomes necessary to the antenna. For a given purpose
For which nature
allows a large number of solutions and indeed quite a r’ew different matching circuits can be found in the literature: a shunt inductance in series with a large capacitor or a A/4 open line for d.c. insulation, an open stub plus a 1/4 transformer, a stepped transmission line transformer. It is beneficial to choose a solution with large bandwidth and a large error budget for the junction capacitance because this value cannot be controlled well by photolithography, once the junction area approaches values as small as 1 x 1 pm. The authors rely on an “endloaded transmission line”‘2’) for matching which is deposited on the antenna patches of the planar dipole (Fig. 9). The superconducting stripline, connecting the 2 junction array, has two parameters, length and width, which can be chosen. The length, measured from the vertex of the dipole to the junction,
longer than a quarter
of a wavelength.
It turns the phase angle of the signal such that the junction, which is essentially shorted by its intrinsic capacitance, is driven by the required conjugate complex impedance. The width of the metalization defines the characteristic impedance of the stripline. It must be chosen such that the desired transformation ratio is accomplished. A wide stripline with a characteristic impedance < 5Q is required e.g. to match a large area junction to the antenna impedance of w 5On. The calculation of length and width is trivial using the theory of low loss transmission lines. It will be discussed below that the stripline must have very small attenuation. 5. Receitler noise For noise calibration all noise sources in the block diagram (Fig. 12) are thought to be replaced by an equivalent source at the input. The noise temperature of this source is calculated from a measurement of IF-noise power for two levels of additional input noise from a calibrated source of noise placed at the receiver entrance. For the actual measurement a thermal power meter is connected to the IF output and absorbers, one at ambient temperature, the other at 77 K are placed
z 8 -J
: Z, = 5,OE+OO
Frequency (20.00 GHz/div.) 1 L = 1.6E-02 1 L, = 2.OE-04
Fig. 10. Calculated frequency response (insertion loss in dB) of a SIS mixer using a matching circuit according to Fig. 9 in order to match a single junction R,= 13 R C, = 12OW into a monpole of 25 R impedance (solid line). This assumption corresponds to a series array of 2 junctions connected to a 50Cl dipole. The additional curves show the error budget: dashed line for 160 fF, dashed dotted line for 90 fF junction
in front of the receiver. Figure 11 shows an example. The noise contribution of the IF amplifier is not constant all over the IF bandwidth. A tunable bandpass filter of 50 MHz bandwidth in between the IF-output and the thermal power meter allows measurement of receiver noise vs IF-frequency. The receiver noise in Fig. 11 is 10 K less than listed in Table 2 because the mixer was not yet collimated
for the operation
of the telescope.
6. Noise budget For better
(Fig. 12). It is necessary
and in view of future
to estimate Taad 4.2K
a noise budget
can be made
of each component
in the signal
0.8 mW 0.6 0.4 0.2
Op 0.8 mW 0.6 0.4 0.2
Fig 11. Noise calibration. The upper measurement is made at 4.2 K, the lower at 1.5 K bath temperature. The curves in the lower right corners show the IF power vs the bias voltage. The symmetrical curves show the bias current versus bias voltage with the LO switched on and off. The measured receiver noise is 80 and 75 K at 4.2 and 1.5 K bath temperature respectively.
Antenna-coupled superconducting contacts
path and to calculate the noise introduced in different stages. The optical transmissions are calculated using Fresnel’s equations. The emissivity follows from the total transmission of the warm optics. The beam efficiency is our best estimate for a dipol-lens combination assuming that the radiation directed towards the rear side is fully recovered by the reflecting backplane. If the photon noise is referred to the mixer input, it becomes independent of the LO power and the corresponding mixer gain. Our value assumes that shot noise, produced by LO induced current, is uncorrelated in a series array of two junctions. The mixer gain is calculated with the given formula which follows from photomixer theory, the validity of which can be shown by expanding the Besselfunctions in the Tucker formulae for low LO levels. (**Jo)The reason for not using the precise theory is that short and explicit expressions can be used for a discussion which is based on an estimated value for the beam efficiency anyway. The noise sources behind the mixer are easily determined. The amplifier added noise was taken from a test report, the bath temperature from the vapor pressure table and shot noise by the dark current is calculated with the given formula. The IF-noise from the amplifier was verified by a shot noise calibration, plotting the increase of IF-noise vs bias current in the normal conducting branch of the I/V curve. It is important to note that the mixer has moderate gain rather than loss in the noise budget. The IF-impedance of the mixer corresponds to the tangential resistance of the pumped I/V curve (Fig. ll), -4OOQ at a bias voltage of 4 mV at the array. With the impedance of the IF-amplifier transformed up to 200 R by a quarter wave line transformer, the mixer works into a load resistance of N 135n which results in a gain of +0.7 dB for a LO-induced current of 10 ,uA. With a moderate mixer gain the photon noise at the mixer input becomes the dominant noise source. Tests by e.g. lowering the l-He bath temperature support the picture of a photon noise limited mixer. The mathematical model (Fig. 12) predicts the observed receiver noise (Fig. 11) reasonably well, however, with our present knowledge of beam efficiency we cannot claim that it is verified by experiment. If our understanding is correct, a lower receiver noise can be expected with a single junction. The photon noise of 16.6 K for an array of two junctions should decrease to 8.3 K. The experiment has not yet been made because the deposition process must be modified. Furthermore the improvement in receiver noise will be not so marked because at lower photon noise amplifier noise will start to dominate the overall performance again. 3.6dB -0.ldB in
Lens Surf. Ref.
Vpe k n,q eioR/k
Fig. 12. Block diagram
Bath [email protected]
cold optics input
n s G 2 forDSB n, = 1 for SSB ni = number of junctions in array vp = nj h v I e plateau voltage p dark current iD iL = LO ind. current q g quantum efficiency of the receiver
and noise budget.
H. ROTHERMELet al.
Fig. 13. Surface resistance
of thin films at 4.2 K: left curve aluminum, middle niobium, right niobiumnitride. The dashed curve indicates an upper limit of surface resistance for a resonant stripline circuit matching a [email protected]
junction, the dashed dotted line for matching a NbAl-Alox-Nb junction.
As discussed above, the operating frequency of a SIS junction can be raised far above the RC limited corner frequency by a resonant matching circuit. The tuning range at the higher operating frequency remains restricted to a value equal to or smaller than the corner frequency. If we call the ratio of operating frequency over the tuning range “loaded quality” we must make sure that the “unloaded quality” of the matching circuit, which is the resonator quality of the stripline, the input and output loads disconnected, is 10 times larger. If not, the insertion loss of the matching circuit will degrade the receiver noise temperature seriously. Figure 13 shows the surface resistance versus frequency for 3 materials at 4 K and 2 types of junctions. The curve for aluminum considers
PA 160 120 80 40 0
0.75 0.5 LOOff I
Fig. 14. Mixing experiment at 216 GHz with a series array of two NbMgO-Nb junctions with areas of 1.6 x 1.8 ym each. The upper curves are the I/V characteristics with the local oscillator on and off. The curves below show the IF-power vs bias voltage with a 290 and a 77 K load at the receiver input. The corresponding DSB receiver noise temperature is 320 K.
skin effect. As long as a material
the surface resistance
with the quality requirement. On the other hand normal conducting Nb, NbN and Al are out of question for the matching circuit. The NbN-MgO-NbN junction has a lower corner frequency due to a high specific capacitance. Its quality requirement cannot be met beyond 600 GHz by superconducting NbN films. Providing NbAl-Alox-Nb junctions work up to twice the gap frequency(24) this type of junction will reach the highest frequency (1060 GHz) with NbN wiring, but this arrangement needs additional research about losses in the contact areas of the junction itself which consist of pure Niobium Niobiumnitride.(25,26)
above gap frequency)
In spite of higher transition temperature and gap frequency Niobiumnitride has not yet relieved the frequency limitation of SIS mixers. Nevertheless it is interesting to do mixer experiments with this material. NbN-MgO-NbN junctions were produced(27) using the mask set for a 345 GHz Nb junction with matching circuit (Fig. 9). The phase velocity in a NbN-SiO,-Nb stripline is lower and the capacitance of the junction is larger. Therefore the response was week at 345 GHz. At 216 GHz a DSB noise temperature of 320 K was found (Fig. 14). Better numbers are expected with a mask set dedicated to the material parameters of NbN. The comparatively soft I/V curve is promising having (Fig. 14) sacrifies quantum efficiency according to theory. (22)The measurement in mind that niobium junctions did not approach the performance of classic PbBi junctions, until efficient matching structures had been worked out. Similar mixing experiments with NbN junctions have been made already.‘2*~29’
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