Dual-layer and dual-polarized metamaterial inspired antenna using circular-complementary split ring resonator mushroom and metasurface for wireless applications

Dual-layer and dual-polarized metamaterial inspired antenna using circular-complementary split ring resonator mushroom and metasurface for wireless applications

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Journal Pre-proofs Regular paper dual-layer and Dual-polarized Metamaterial Inspired Antenna using CircularComplementary Split Ring Resonator Mushroom and Metasurface for Wireless Applications Mohammad Ameen, Raghvendra Kumar Chaudhary PII: DOI: Reference:

S1434-8411(19)31716-9 https://doi.org/10.1016/j.aeue.2019.152977 AEUE 152977

To appear in:

International Journal of Electronics and Communications

Received Date: Revised Date: Accepted Date:

10 July 2019 20 October 2019 28 October 2019

Please cite this article as: M. Ameen, R. Kumar Chaudhary, dual-layer and Dual-polarized Metamaterial Inspired Antenna using Circular-Complementary Split Ring Resonator Mushroom and Metasurface for Wireless Applications, International Journal of Electronics and Communications (2019), doi: https://doi.org/10.1016/ j.aeue.2019.152977

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Dual-layer and Dual-polarized Metamaterial Inspired Antenna using Circular-Complementary Split Ring Resonator Mushroom and Metasurface for Wireless Applications Mohammad Ameen1, and Raghvendra Kumar Chaudhary2 Department of Electronics Engineering, Indian Institute of Technology (Indian School of Mines) Dhanbad, Jharkhand-826004, India Email: [email protected]; [email protected] Corresponding author:- Raghvendra Kumar Chaudhary, [email protected] Abstract—A compact triple-band and dual-polarized metamaterial inspired antenna loaded with a reactive impedance surface (RIS) based metasurface is presented in this work. The dual-band circular polarization (CP) characteristics are obtained by the combination of the conventional slotted patch loaded with circular-complementary split-ring resonator based mushroom stacked with RIS. The CP radiation is generated due to the introduction of asymmetry in the antenna structure results in the splitting of fundamental mode into two orthogonally polarized modes with 900 phase difference at the resonating frequency. The antenna obtains an overall electrical size of 0.27λ0 × 0.27 λ0 × 0.027λ0 with radiating element size of 0.16λ0 × 0.16 λ0 at 1.7 GHz with ka = 0.72. The final antenna prototype is fabricated and measured which covers bandwidths of (1.66–1.72 GHz) 3.52%, (1.94–2.20 GHz) 13% and (3.68–3.87 GHz) 5.02% for the three bands. The antenna exhibits CP radiation at 2.10 GHz and 3.78 GHz with an axial ratio bandwidth (ARBW) of 2.85% and 1.85% are obtained for the two bands, respectively. Due to the loading of RIS, the antenna gain enhanced to 1.6 dBi, 4.74 dBic, and 3.28 dBic at the three respective frequencies with an overall area reduction of 52.32% is achieved. The antenna provides a wider AR beamwidth for CP radiation (ARBW<3) with positive gain values for theta = ±580 and for all phi values at 2.1 GHz. Hence, the proposed antenna can be used for working in 1.7 GHz LTE (1.66–1.72 GHz), 2.2 GHz UMTS (1.94–2.20 GHz), 3.8 GHz WiMAX (3.68–3.87 GHz), and small satellite application systems at 2.1 GHz applications. 1

Keywords— circular polarization; co-axial feeding; complementary split-ring resonator; metasurface; reactive impedance surface; triple band.

I. INTRODUCTION Multi-functionality with miniaturized low profile wireless gadgets and associated components are needed for the upcoming wireless systems. In the modernized world of emerging applications, antenna engineers and industries are working towards device miniaturization with a lower cost of fabrication. The wireless communication systems prefers the use of microstrip antennas with multiple functionality that can simultaneously support devices operating at various frequency bands such as 4G-long term evolution (LTE) (1.71–1.88 GHz, 2.3–2.4 GHz), worldwide interoperability for microwave access (WiMAX) (3.3–3.4 GHz, 3.5–3.8 GHz), and universal mobile telecommunication services (UMTS) (1.88–2.2 GHz, 2.5–2.69 GHz) due to the characteristics of lightweight, low profile, ease of integration with other devices, and easy fabrication process. Unfortunately, due to technology advancements towards antenna miniaturization and performance enhancement, the existing patch antennas are not suitable to perform what the current technology needed. It mainly faces the problem of narrow bandwidth, poor gain, larger antenna size, higher level of cross-polarization, and smaller radiation efficiency [1]. The concept of metamaterials (MTMs) is introduced by Smith. et al. [2], after that progressively new antenna designs are emerging with amazing changes in the antenna industry. MTMs are artificial homogeneous electromagnetic structures having unusual properties of negative propagation constant (β <0), negative permittivity (ε<0) or permeability (μ<0). Those with μ < 0 are represented as a munegative (MNG) MTM [3] and ɛ < 0 are represented as epsilon-negative (ENG) MTM. The split ring resonators (SRR) fall in the category of MNG-MTMs and complementary split-ring resonator (CSRR) belongs to ENG-MTMs [3, 4], which can be used to design MTM inspired antenna with miniaturized size. These antennas use composite right/left-handed (CRLH) transmission line (TL) [1], ENG-TL radiators [5], MNG-TL radiators [6], resonant approaches using MTM inspired SRR [7, 8], CSRR loadings [9], and Koch fractal MTM based electric field-driven LC resonator loading [10]. Recently, the usage of circularly polarized (CP) antennas are acquiring more consideration among small satellite communication systems and various applications due to insensitivity towards the placement of antenna, better immunity towards faraday rotation effects caused by the ionosphere, 2

minimizes multipath effects and allows more flexibility in the placement of transmitter and receiver antenna [5]. Various number of MTM based CP antennas are reported in [6], [11]–[13]. These antenna uses rectangular radiators with chip capacitor and vias for realizing MNG MTMs [6], dual-polarized antenna utilizing CRLH-TL unit cell with CSRR for miniaturization [11], patch antenna loaded with CSRR mushroom unit cell loadings [12] obtains a higher size of 60 × 60 × 1.6 mm3, and trimmed patch antenna utilizing (+1 mode) and CRLH-TL based triangular mushrooms utilizing (-1 mode) for dual CP radiation are discussed in [13]. These antennas provide an excellent level of compactness but fail to provide minimum bandwidth needed for various wireless applications. Also, most of the designs, gain tends to be very less or negative and due to this radiation efficiency is very small. Such antennas are unsuitable or fewer chances for use in modern application systems. To alleviate the above explained-problems the loading metasurfaces (MS) such as artificial magnetic conductors (AMC), reactive impedance surface (RIS), electromagnetic band-gap (EBG) and high impedance surfaces (HIS) combined with the radiator antenna, the radiation characteristics can be improved with compact size. The antennas use a pair of square-shaped CSRR mushroom based metaresonator combined with RIS for single band and multiband antenna performances with different polarization states are explained [9], the combination of Wunderlich-shaped fractal CSRR loaded on the radiator patch and Hilbert shaped RIS loaded in the ground plane for miniaturization [14], the combination of complementary crossbar fractal resonator and three-turn complementary spiral resonators combined with RIS for single and multiple band CP antenna applications [15], dual-mode dual-polarized antenna using crossed dipoles with AMC obtains a large antenna size of 125 × 125 × 6.50 mm3 [16], symmetrical bowtie shaped patches loaded with U shaped slot radiator combined with AMC constitutes a size of 104 × 104 × 11 mm3 [17], square patch stacked with modified mushroom unit cell loaded EBG obtains a size of 70 × 70 × 2.5 mm3 [18], stacked CP antenna using concentric annular ring patches with a larger size of 140 × 140 × 12 mm3 are explained in [19], truncated monopole antenna stacked with 2×2 CSRR loaded MS [20], annular slot radiator with HIS ground plane by combining the microstrip mode and annular slot mode with higher antenna volume of 150 × 150 × 18 mm3 [21]. Wideband CP antenna using a square patch radiator and AMC reflector [22], trimmed patch antenna with square MS [23], and spiral slots with MS acquire a larger volume of 93.29 × 93.29 × 27

3

mm3 [24]. These explained antennas provide good radiation characteristics, but the overall sizes are too large and bulky due to the higher antenna profile. In this work, a compact multi-band and dual-polarized antenna is designed to reduce the drawbacks suffered by the antennas interpreted in [11]–[23], where these antennas show larger dimensions, narrow impedance bandwidths, narrow ARBW, poor gain, and lower radiation efficiency. The proposed antenna obtains a comparable size of 0.27λ0 × 0.27λ0 × 0.027λ0 with an area reduction of approximately 52.32% is achieved when compared with conventional patch antenna. All the simulation and analysis are done using the CST Microwave Studio suite. The proposed antenna design is divided into six sections. Firstly, a multi-band CP antenna is designed by loading of circular-shaped CSRR metaresonator in a conventional patch is explained in section II. Section III describes the design and analysis of 5×5 unit cell loaded RIS forming a MS. In section IV, combining the antenna designed in section II with MS designed in section III for obtaining compactness, multi-band CP, good gain, and acceptable radiation efficiency. Section V explains the experimental results and comparison with existing designs. Section VI explains the conclusion of the proposed work. The proposed antenna finds suitable applications in 1.7 GHz LTE (1.66–1.72 GHz), 2.2 GHz UMTS (1.94–2.20 GHz), 3.8 GHz WiMAX (3.68–3.87 GHz), and small satellite applications at 2.1 GHz.

II. ANTENNA GEOMETRY AND DESIGN C-CSRR loaded Mushroom Coaxial feeding Patch antenna with C-CSRR loaded mushroom

Z H1 H2 Y

Top Radiator Layer Bottom RIS Layer

X

SMA connector Via

Z RIS unit cells

(a)

X

Y (b)

Fig. 1. Triple-band dual-layer and dual-polarized MTM inspired antenna. (a) Two-dimensional front view, and (b) Threedimensional top view.

The three-dimensional configuration of the MTM inspired antenna loaded with RIS is depicted in Fig. 1(a), which primarily consists of a top dual-polarized multi-band antenna of thickness H1 which is stacked by RIS based MS of thickness H2. Fig. 1(b) depicts the three-dimensional, top view of the final RIS loaded MTM inspired dual-polarized antenna. All the design stages will be clearly explained in detail in the upcoming stages. 4

A. Design Stages of the Proposed Metamaterial Inspired Antenna Square Patch with slot

Square Patch

C-CSRR loaded Modified Mushroom

RIS Loaded Antenna

Feed point (Xf,Yf)

L1

a W1 Antenna 1

Antenna 2

Antenna 3

Antenna 4

(b)

(c)

(d)

(a)

Fig. 2. Design stages of the proposed MTM inspired antenna. (a) microstrip square patch antenna, (b) square patch antenna with a square slot, (c) C-CSRR mushroom unit cell loaded antenna, and (d) MTM inspired antenna loaded with RIS. 0 -5

Axial Ratio (dB)

12

S11(dB)

-10 -15 -20 Antenna 1 Antenna 2 Antenna 3 Antenna 4

-25 -30 -35 1.5

2.0

2.5

3.0

3.5

4.0

Antenna 2 Antenna 3 Antenna 4

9 6 3 0

4.5

5.0

Frequency (GHz)

2.0

2.1

2.2

2.3

3.7

3.8

3.9

4.0

Frequency (GHz)

(a)

(b)

Fig. 3. proposed antenna design stage responses from Antenna-1 to Antenna-4 (a) input reflection coefficient (S11), and (b) ARBW.

Fig. 2(a) to 2(d) represents the design stages of the proposed MTM antenna from Antenna-1 to Antenna-4. The input reflection coefficient and ARBW responses for various design stages are illustrated in Fig. 3(a) and 3(b) respectively. Firstly, a conventional square-shaped patch antenna is designed as described in Fig. 2(a) represented by Antenna-1. Then a square-shaped slot added on the top right corner for the generation of another mode shown in Fig. 2(b) represented as Antenna-2. Fig. 2(c) illustrates the loading of C-CSRR loaded mushroom metaresonator denoted by Antenna-3 and finally the RIS based MS is stacked with the radiator antenna shown in Fig. 2(d) denoted by Antenna4. Table. I illustrate the performance characteristics of the antenna at various design stages (Antenna-1 to Antenna-4) regarding the number of operating bands, antenna electrical size, ka value, impedance bandwidth, ARBW and reason for various bands.

5

TABLE I. COMPARISON OF THE SIMULATED ANTENNA PERFORMANCE FROM ANTENNA-1 TO ANTENNA-4 Radiating No: of Freq. Ka* Imp. ARBW Antenna element size Reason for the Bands Bands (GHz) value BW (%) (%) (λ03) 2.41 – – Due to square patch 0.23 × 0.23 × Antenna-1 2 1.03 0.012 4.81 6.97 – Higher-order mode 2.20 3.09 – Due to square patch 0.21 × 0.21 × Antenna-2 3 2.39 0.94 2.92 – Rectangular slot loading 0.011 4.48 2.67 – Due to Higher-order mode Merging of resonances due 2.31 6.14 1.35 to metaresonator and 0.22 × 0.22 × Antenna-3 2 0.99 slotted patch 0.011 4.46 2.35 – Due to Higher-order mode 1.66 4.27 – Due to metaresonator Merging of resonances due 2.05 0.16 × 0.16 × 14.63 2.36 to square patch and Antenna-4 3 0.71 0.02 rectangular slot Higher-order modes and 3.78 3.96 2.09 RIS metasurface *

Note: In all the antenna design stages, the ka value is calculated on the basis of the first resonating frequency. Also in Antenna-4, the final resonance at 4.55 GHz is not considered.

B. Conventional Square Microstrip Antenna Design A conventional square-shaped patch antenna is considered as the reference antenna [25] as demonstrated in Fig. 2(a) which operates around 2.5 GHz. The antenna is designed on FR-4 substrate with relative permittivity (εr) of 4.4, loss tangent (tan δ) of 0.02 with thickness H1 = 1.6 mm. Coaxial feeding scheme is used in the design with feed point located at (Xf = -13 mm, Yf = 0 mm) with patch width W1 = 29 mm, and length L1 = 29 mm obtains a total radiating element size of 29 mm × 29 mm (0.24 λ0 × 0.24 λ0) at 2.5 GHz and ground plane dimensions of 48 mm × 48 mm × 1.6 mm (0.40 λ0 × 0.40 λ0 × 0.013 λ0) at 2.5 GHz. C. Design of C-CSRR Loaded Modified Mushroom Unit cell W4

Conventional square unit cell L4

Conventional mushroom Via r = 0.3 mm

R2

R1

G3

(Xl, Yl) G2

Top view

G1

(a)

H1

Via

Side view (b)

(c)

Fig. 4. Design of C-CSRR loaded mushroom unit cell (a) simple square patch, (b) conventional mushroom unit cell, and (c) MTM inspired C-CSRR loaded mushroom unit cell.

6

Fig. 4(a) to Fig. 4(c) illustrates the schematic of top view and side view for the design stages of C-CSRR loaded mushroom. At first, a square unit cell of dimension W4 × L4 is designed as depicted in Fig. 4(a). Further achieving compactness, a metal post or via of radius r = 0.4 mm is inserted in the middle of square unit cell as shown in Fig. 4(b). It is characterized by parallel a LC resonator [26]. For further reducing the resonant frequency without increasing unit cell size, a C-CSRR slot is inserted into the top side of the mushroom unit cell as depicted in Fig. 4(c). The resonance frequency of C-CSRR can be written as eqn. (1) 𝑓𝑐𝑠𝑟𝑟 =

1 2𝜋 √𝐿𝑒𝑞_𝑐𝑠𝑟𝑟 𝐶𝑐𝑠𝑟𝑟

(1)

where 𝐿𝑒𝑞_𝑐𝑠𝑟𝑟 = 𝐿𝑐𝑠𝑟𝑟 ⁄4 and 𝐿𝑐𝑠𝑟𝑟 = 2𝜋𝑅2 𝐿𝑝𝑢𝑙 , where 𝐿𝑝𝑢𝑙 denotes the inductance per unit length of CPW connecting the inner circular disc to ground [27]. The capacitor 𝐶𝑐𝑠𝑟𝑟 is represented by the metallic disc of radius 𝑅2 − (𝐺2 ⁄2) encircled by the ground plane at a length 𝐺3 of its edge [27]. Thus the CSRR resonant frequency (𝑓𝑐𝑠𝑟𝑟 ) can be reduced by increasing the capacitance (Ccsrr) or inductance (Lcsrr), ie, the square patch width (W4) and gap G2, dielectric permittivity and substrate thickness (H1). Thus the C-CSRR significantly increases the capacitance LC circuit and hence the resonant frequency decreases. The new resonances are generated [26] due to the capacitance generated by the voltage gradients between CSRR gaps (G2) and the inductances generated by the currents flowing through C-CSRR coils (with width G1 and G3), and their connections with the via. Thus a compact multiband band unit cell can be designed. The C-CSRR loaded mushroom unit cell illustrated in Fig. 4(c) designed using low cost FR-4 substrate of εr = 4.4 and tanδ = 0.02 with a height H1 = 1.6 mm. The optimized dimensions are W4 = L4 = 8 mm, R1 = 3.5 mm, R2 = 2.5 mm, G2 = G1 = 0.4 mm, and G3 = 0.5 mm, r1 = 0.3 mm and (Xl, Yl) = (7.2, 7.2). The overall dimensions of the C-CSRR mushroom unit cell are 8 mm × 8 mm× 1.6 mm (0.066 λ0 × 0.066 λ0 × 0.013 λ0 at 2.5 GHz). The implementation of metaresonator inside the antenna plays a substantial role in the performance characteristics of the antenna depicted in Fig. 1. Therefore it is necessary to investigate the characteristics of metaresonator at the resonance frequencies. Consider Fig. 5(a) in which the metaresonator is terminated with transmission lines on both sides and the corresponding equivalent circuit diagram is shown. The transmission (S21) and reflection (S11) coefficients are extracted using CST Microwave Studio using the setup depicted in Fig. 5(b) are plotted in Fig. 5(c). The transmission and reflection coefficient predicts a passband filter characteristics two the frequencies at 2.25 GHz and 7

Port-2

CSRR

Transmission Line

Microstrip feed via

Transmission Line

Ccsrr

L1

L2

Via

Lcsrr/2 Lcsrr/2

Lvia

C1

C2

Port-1

CSRR Mushroom unit cell

-2 Port

-1 Port Z

Y X

0 -5 -10 -15

First order

-20 Second order Simulated S11 Simulated S12 Equivalent ckt. S11 Equivalent ckt S12

-25 -30 -35 -40

Ground plane

1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5

FR-4 substrate (b)

Frequency (GHz)

(c)

Conventional mushroom unit cell

4.5 4.0

Conventional case CSRR Mushroom

3.5

RH Region

3.0 CSRR Mushroom unit cell

Permittivity (ε )

Frequency (GHz)

5.0

2.5 LH Region

2.0 0

20 40 60 80 100 120 140 160 180

150 120 90 60 30 Real (ε) 0 -30 -60 -90 -120 1.5 1.8

d (degrees)

(d)

Real (ε)

5 4 3 2 1 0 -1 -2 -3 -4

Real (μ)

Real (ε) Imag (ε) Real (μ) Imag (μ)

2.1 2.4 2.7 3.0 Frequency (GHz)

3.3

Permeability (μ)

Microstrip feed

Scattering Parameters (dB)

(a) via CSRR

3.6

(e)

Fig. 5. C-CSRR loaded mushroom characteristics. (a) CSRR loaded mushroom with microstrip line feed and corresponding equivalent circuit diagram, (b) Two-port set up for the analysis of CSRR mushroom, (c) Transmission and reflection coefficients, (d) dispersion diagram plots of the proposed and conventional unit cell, and (e) real and imaginary permittivity response.

4.1 GHz. Using the equivalent circuit mention in Fig. 5(a) the circuit simulation is done using Keysight ADS software. The tuned values of inductors and capacitors are L1 = L2 = 6.3 nH, C1 = C2= 0.397 pF, Lcsrr1 = Lcsrr2 = 3.9 nH, Ccsrr = 3.08 pF, and Lvia = 3.782 nH. Also, the two passbands satisfy the equivalent circuit simulation and 𝑓𝑐𝑠𝑟𝑟 calculated using eqn. (1) as 𝑓𝑐𝑠𝑟𝑟 =

1 2𝜋 √𝐿𝑒𝑞_𝑐𝑠𝑟𝑟 𝐶𝑐𝑠𝑟𝑟

= 2.08 GHz. Thus, it

assigns to the first order passband, defined to be 𝑓𝑐𝑠𝑟𝑟 Similarly, the second-order passband is achieved at 2 × 𝑓𝑐𝑠𝑟𝑟 = 4.16 GHz depicted in [19]. Thus, it is the evidence of CSRR loaded mushroom contributes to achieving multiband antenna performance. Fig. 5(c) shows the comparison between dispersion plots 8

for conventional mushroom and C-CSRR loaded mushroom based on the CST simulated scattering parameter results plotted using eqn. (2) in ref. [1] 1 − 𝑆11 𝑆22 + 𝑆12 𝑆21 𝛽𝑑 = 𝑐𝑜𝑠 −1 ( ) 2 × 𝑆21

(2)

It can be noted that the conventional mushroom will work at nearly 3.75 GHz and whereas the proposed C-CSRR mushroom provides zero phases nearly at 2.5 GHz and 4.0 GHz confirms MTM property [3]. Comparing with Fig. 5(c) and Fig. 5(d), there is a slight shift in the passband frequencies. It is concluded that the unit cell can provide two passbands (S21 = 0) nearly at 2.25 and 4.0 GHz respectively. Also, the proposed C-CSRR loaded unit cell achieves a 44% electrical size reduction compared with the conventional mushroom unit cell (Fig. 4(b)) at the same frequency. The material parameters of the C-CSRR mushroom metaresonator are calculated using the extracted scattering parameters using the set up mentioned in Fig. 5(b). The permittivity (ε) and permeability (μ) characteristics of the C-CSRR unit cell are extracted using the method explained by Chen et. al [28] where the permittivity is given by Eqn. (3) and (4) 𝜀=

𝑛 𝑧

(3)

𝜇 = 𝑛𝑧 where 𝑛 is the refractive index, 𝑛 = 𝑒 𝑖𝑛𝑘0 𝑑 =

𝑆21

𝑧−1 1−𝑆11 𝑧+1

′′

1 𝑘0

(4) ′

[[ln(𝑒 𝑖𝑛𝑘0 𝑑 )] − 𝑖[ln(𝑒 𝑖𝑛𝑘0 𝑑 )] ], the exponential term 𝑑

′′



, [ln(𝑒 𝑖𝑛𝑘0 𝑑 )] denotes the complex component, [ln(𝑒 𝑖𝑛𝑘0 𝑑 )] denotes the real

component of the complex number, 𝑘0 is the wavenumber, 𝑑 is the maximum length of the unit element. (1+𝑆 )2 −𝑆 2

The impedance is given by 𝑧 = ±√(1−𝑆11 )2 −𝑆21 2 , 𝑆11 and 𝑆21 are the reflection and transmission 11

21

coefficients respectively. The extracted real and imaginary permittivity responses of the C-CSRR unit cell are displayed in Fig. 5(e). It exhibits negative values of permittivity and positive permeability at 2.25 GHz due to a first-order passband of the C-CSRR as depicted in Fig. 5(c). Also, from Fig. 5(e), a peak value of 145 in the extracted real part of permeability corresponding to the magnetic resonance of the metaresonator. The extracted material parameter graph in Fig. 5(e) reveals that the CSRR loaded mushroom unit cell is based on ENG material in the frequency range of 2.25 GHz. Hence, it confirms the existence of MTM behavior [3]. 9

4.2

4.3

4.1

4.2

4.0

Frequency (GHz)

Frequency (GHz)

D. Parametric Analysis of the proposed C-CSRR Mushroom Unit Cell

Second mode First mode

2.3 2.2 2.1 2.0

4.1 2.4

Second mode First mode

2.3 2.2 2.1

1.9 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1

2.0 2.2 2.4 2.6 2.8 3.0 3.2 3.4 3.6 3.8

Variation of r (mm)

Variation of R1 (mm)

(a)

(b)

Fig. 6. C-CSRR mushroom unit cell characteristics. (a) variation of via radius (r), and (b) variation of CSRR outer ring radius (R1).

To analyze the performance of the metaresonator, a parametric study is conducted by varying the dimensions of the via radius (r) and ring radius (R1) as pictured in Fig. 6(a) and 6(b) respectively. From Fig. 5(c), it is identified that the proposed mushroom unit cell can generate two passbands at 2.25 GHz and 4.1 GHz. From Fig. 6(a), it identifies that the change in r can vary the first pass passband linearly. But a change in r will not have any effect on the second passband. Also, Fig. 6(b) the variation of CSRR outer radius R1 identifies that increasing the radius R1 shifts the two passbands to lower frequency. After optimization an optimum value of r = 0.4 mm and R1 = 3.5 mm is chosen for fabrication. E. Design of Dual-Polarized MTM Inspired Antenna Via

W4

a L2 L

L1

W2 L3 Probe feed

W Z

X

R1

L4

(Xf,Yf)

W1

Y

R2

(Xl, Yl) G2 Patch

G1 C-CSRR Copper Substrate

(a)

(b)

Fig. 7. Multiband MTM inspired antenna (a) schematic view of the multiband MTM inspired antenna with an enlarged view of C-CSRR mushroom unit cell, and (b) fabricated antenna prototype.

The diagrammatic configuration of the proposed dual-polarized dual-band MTM inspired CP antenna is depicted in Fig. 7(a). Initially, a square patch antenna is designed as depicted in Fig. 2(a). Then a square slot of size W2 × L2 is cut from the patch at top right corner. The slotted square patch 10

antenna generates two modes at 2.2 GHz and 2.3 GHz. Also at 4.5 GHz another band obtained depicted as Antenna-2 [See Fig. 3(a)]. Then a C-CSRR mushroom metaresonator is added in the square slot for further size reduction which will combine the two orthogonal modes (2.2 GHz and 2.3 GHz) and generates CP radiation at 2.25 GHz depicted as Antenna-3 [see Fig. 3(a) and 3(b)] respectively. Also, second resonance at 4.5 GHz remains unchanged and hence a dual-band dual-polarized MTM inspired antenna is designed. Fig. 7(b) depicts the fabricated antenna prototype (Antenna-3). The overall size of the antenna is 0.35 λ0 × 0.35 λ0 × 0.011 λ0 at 2.2 GHz and radiating element size of 0.21 λ0 × 0.21 λ0 at 2.2 GHz. The optimized antenna dimensions are L = 48 mm, L1= 29 mm, L2 = 11.2 mm, L3 = 16.6 mm, L4 = 8 mm, W = 48 mm, W1 = 29 mm, W2 = 11.2 mm, W4 = 8 mm, R1 = 3.5 mm, R2 = 2.5 mm, G1 = G2 = 0.4 mm, and a = 1.4 mm. F. Circular Polarization Mechanism of Antenna-3 The studies on the CP antenna demonstrate that by introducing asymmetry [29] in the design can generate CP radiation due to the splitting of fundamental modes into two degenerate modes. Hence the perturbation structures implemented on the square patch depicted in Fig. 2(a) can generate orthogonally polarized modes with 900 phase difference. Here in the proposed antenna, the slot strategy for CP generation is utilized [see Fig. 2(b)]. Using this idea a C-CSRR loaded mushroom unit cell (metaresonator designed in section II. c) is loaded in the patch antenna depicted in Fig. 2(c) is responsible for generating CP waves and facilitate antenna miniaturization [29]. Due to the asymmetric structure, the radiated beam of the antenna is left-handed CP (LHCP) or right-handed CP (RHCP) depending on the placement of the slot and metaresonator inside the square patch as depicted in Fig. 8(a)–8(d). Fig. 8 depicts the study on the placement of metaresonator on different quadrants. The unit cell is placed in the quadrant-2 (Region-1) and quadrant-3 (Region-2) near to the probe feed depicted in Fig. 8(a) and 8(b) can generate RHCP and LHCP radiation respectively. The corresponding S11 responses and ARBW responses are plotted in Fig. 9(a) and 9(b). Similarly, Fig. 8(c) and 8(d) depict the placement of the unit cell in quadrant-4 (Region-3) and quadrant-1 (Region-4) can generate RHCP and LHCP radiations respectively. Fig. 9(a) and 9(b) depict the S11 and ARBW responses due to the placement of metaresonator in the region-3 and region-4 respectively. Here, the better CP radiation at 2.22 GHz with good impedance matching is obtained for the metaresonator placement at Region-3 (RHCP) and Region-4 (LHCP). For small satellite antenna applications, LHCP antennas are more 11

useful. Hence it can be concluded that the antenna depicted in Fig. 8(d) is more useful in small satellite applications. Region-1

Square Patch

Region-4 Metaresonator

Feed point

Region-3

Region-2 (a)

(b)

(c)

(d)

Fig. 8. Studies on the placement of C-CSRR mushroom unit cell in different quadrant. (a) quadrant-2 (Region-1), (b) quadrant-3 (Region-2), (c) quadrant-4 (Region-3), and (d) quadrant-1 (Region-4). 0

14

Axial Ratio (dB)

-5

S11(dB)

-10 -15 -20

Region 1 Region 2 Region 3 Region 4

-25 -30 2.2

2.4

2.6 4.0

4.2

4.4

12 10 8 6 4 2

-35 2.0

Region 1 (RHCP) Region 2 (LHCP) Region 3 (RHCP) Region 4 (LHCP)

4.6

4.8

0 2.10

2.15

2.20

2.25

2.30

2.35

Frequency (GHz)

Frequency (GHz)

(a)

(b)

Fig. 9. Simulation results on the placement of CSRR loaded mushroom unit cells in different quadrant. (a) Input reflection coefficient (S11) and (b) ARBW responses.

The CP principle explained here is based on the overlapping two nearby orthogonal modes [9] and excite these two modes with a phase difference of 900. Here the probe feed is placed at the edge of the patch [see Fig. 7(a)] along the x-axis and the EM wave proceeds to the two opposite diagonal lines with a phase delay of 450 and automatically 900 phase difference can be introduced. The reason is that at the antenna resonance frequency, the EM wave travels from one edge to the opposite edge with a phase delay of 1800. Now the EM wave propagates from the probe feed to the diagonal line covering an angle of 450, which introduces a phase delay of 450. Also, the impedance matching is improved by varying the position of probe feed and resonance frequency can be controlled by varying the dimension of C-CSRR based metaresonator. It can be observed that when the input signal phases 00 and 1800, the TM01 mode dominates the antenna radiation as depicted in Fig. 10(a) and 10(c). Similarly, when the input signal phases are 900 and 2700, the TM10 mode dominates as shown in Fig. 10(b) and 10(d). Therefore, the two radiation modes can operate independently because they have a phase difference of 900 and hence CP radiation is obtained.

12

ωt = 0°

ωt = 90°

ωt = 180°

ωt = 270°

(a)

(b)

(c)

(d)

Y Z

X

Fig. 10. Simulated surface current distribution of the MTM CP antenna at 2.25 GHz (a) ωt = 0°, (b) ωt = 90°, (c) ωt = 180°, and (d) ωt = 270° (ωt represents the excitation phase).

To examine the CP mechanism at 2.22 GHz, consider the antenna described in Fig. 8(d) with input reflection coefficient response and ARBW in Fig. 9(a) and 9(b) in Region-4. The first and second resonances centered at 2.2 GHz and 2.3 GHz are the two orthogonal modes (TM01 and TM10) of the slotted square patch antenna and hence a phase difference of 900 is introduced between the two modes and generate CP radiation. Fig. 10(a)–10(d) depicts the surface current distribution at 2.25 GHz. Surface current distribution at excitation phase ωt = 0° is due to the current flowing vertically upwards, which is equal in magnitude and opposite direction to that of ωt = 180°. Similarly at ωt = 90° current flowing towards horizontally, which also equal in magnitude and opposite lower left side to that of ωt = 270°. It is true for the case of the designed antenna and hence the criterion for CP is satisfied at 2.25 GHz. The surface current sequentially rotates in a clockwise direction which confirms that the sense of polarization is left-handed CP.

III. DESIGN OF METASURFACE BASED RIS GROUND PLANE For the design of RIS ground plane, 2×2 square patches (unit cell) as shown in Fig. 11(a) is optimized using floquet port boundary condition in CST Microwave Studio to obtain the desired frequency bands between the perfect magnetic conductor (PMC) (0° reflection phase) and perfect electric conductor (PEC) (180° reflection phase) and boundary limits for obtaining the maximum bandwidth with higher level of miniaturization. The corresponding reflection phase diagram is depicted in Fig. 11(b) which operates in 0.6–2.77 GHz frequency range, can cover the first two resonances of the MTM inspired CP antenna (Antenna-3) explained in Fig. 7(a). Also, 1800 reflection phase obtained at 3.95 shows PEC property. The RIS based MS is designed by repeatedly arranging the square unit cells along the x-axis and y-axis to cover the complete ground dimension of La × Lb. The optimized dimensions are La = Lb = 48 mm, Wa = Wb = 7 mm, and G = 0.3 mm. Fig. 11(c) displays the top view and fabricated prototype of the proposed 5 × 5 square unit cell loaded RIS forming a MTM surface for 13

antenna compactness and improve radiation characteristics [30]–[31]. The proposed RIS based MS is fabricated as depicted in Fig. 11(d). Reflection Phase (Degrees)

Waveport Simplified eq. ckt

PMC

PEC PMC

PEC Open

Z

RIS unit cell Y X

200 1800 100 00

0 -100

0.66 - 2.77 GHz RIS operating range

-200 1

Waveport

2

3

4

5

6

Frequency (GHz)

(a) Lb

(b)

Wa G

Wb

Via

Hole for feed line

La

Y

Z

X

(c)

(d)

Fig. 11. RIS design, (a) Metasurface unit cell enclosed by the boundary conditions (floquet port), (b) reflection phase response, (c) schematic of the metasurface, and (d) fabricated MS ground plane.

Cm Lm Cm Lm

Cmg Z1 Cmg

Lm Z2

Cmg

Fig. 12. Equivalent circuit representation of the RIS based MS unit cell depicted in Fig. 11(a).

The proposed RIS can be accomplished by a transmission line model illustrated in Fig. 12 where the capacitance (Cm) is generated due to spacing between the square patches and shunt inductance (Lm) is provided by the square patch of size Wa×Wb, which is parallel with the coupling capacitor (Cmg) generated due to the voltage gradients between the square patch and ground plane. Based on the values of this Lm and Cm, the working frequency of RIS can be written as 𝑓𝑅𝐼𝑆 = 1⁄2𝜋 √𝐿𝑚 𝐶𝑚 , this can be tuned as capacitive or inductive RIS. Thus, its total surface impedance can be represented in as 𝑍(𝜔) = 14

𝑍1 //𝑍2 = 𝑗(1 − 𝜔2 𝐶𝑚𝑔 𝐿𝑚 )⁄𝜔(𝐶𝑚 𝐶𝑚𝑔 𝐿𝑚 𝜔2 − 𝐶𝑚 − 𝐶𝑚𝑔 ). The operating frequency (𝑓𝑅𝐼𝑆 ) of the RIS can be computed using eqn. (4) 𝑓𝑅𝐼𝑆 = 1⁄2𝜋 √(𝐶𝑚 + 𝐶𝑚𝑔 )⁄(𝐶𝑚 𝐶𝑚𝑔 𝐿𝑚 )

(4)

The values of inductance (Lm) and capacitance (Cm) are calculated using eqn. (5) and eqn. (6) explained in [32] 𝑊𝑎

𝐿𝑚 = 2 × 10−4 [ln (𝑊

𝑏 +𝑇

) + 1.193 + 0.2235

𝑊𝑏 +𝑇 𝑊𝑎

] × 𝑘𝑔

(5)

𝑊

𝑤ℎ𝑒𝑟𝑒 𝑘𝑔 = 0.57 − 0.145 ln ( 𝐻 𝑏) 2

𝐶𝑚 = 16.67 × 10−4 𝑊𝑎

√𝜀𝑟𝑒

(6)

𝑍0

where 𝑊𝑎 and 𝑊𝑏 are the width and length of square unit cell, 𝑇 is the thickness of square patch, H2 is the substrate thickness, characteristic impedance (𝑍𝑜 ) and 𝜀𝑟𝑒 is the effective dielectric constant. The LC circuit depicted in Fig. 12 shows inductive nature at frequencies smaller than 𝑓𝑅𝐼𝑆 , open-circuited at 𝑓𝑅𝐼𝑆, and capacitive nature above 𝑓𝑅𝐼𝑆 [30]–[31]. An inductive RIS fixed below the radiator antenna enhances the drawbacks of conventional PEC and PMC surfaces. Firstly, for PEC and PMC surfaces with a real value of impedances, the image currents are focussed on image point and hence it results in a higher value of mutual coupling. For an inductive RIS (reactive impedance η = jv), the image current develops in a sinusoidal manner which is allocated in space, thus reducing the mutual interaction with the source and reduction in the coupling between the radiating antenna and RIS ground plane, which develops a good impedance matching of the antenna with more bandwidth. Secondly, it merges the capacitive behavior of the radiator patch and inductive behaviour of RIS or vice versa results in a smaller frequency than the actual frequency, thus exhibiting antenna overall size reduction, and finally enhances the front-to-back ratio of radiation pattern by reducing the back lobes.

IV. MULTIBAND CIRCULARLY POLARIZED ANTENNA LOADED WITH RIS METASURFACE For the design of proposed dual-layer multi-band and dual-polarized CP antenna, combining the single MTM inspired antenna with RIS based MS are depicted in Fig. 1(a). The top radiating layer is CP antenna loaded with a C-CSRR mushroom-based metaresonator with height H1 = 1.6 mm. Below the radiator antenna, RIS is stacked as a ground plane for the radiating antenna without modifying the size with a thickness of H2 = 3.2 mm. It is observed that the proposed antenna depicted in Fig. 1 is able to provide dual CP bands when loaded with RIS based MS designed in section III. It is evident from 15

Fig. 3(b), the antenna is capable of providing dual CP bands at 2.10 GHz and 3.78 GHz. Also due to RIS loading, the antenna is capable of providing triple-band characteristics with bands centred at 1.66 GHz, 2.10 GHz and 3.78 GHz as depicted in Fig. 3(a). Due to the loading of RIS, the first resonance shifted to the lower frequency at 1.7 GHz. After optimization, the overall dimensions of the antenna are obtained as 0.27 λ0 × 0.27 λ0 × 0.027 λ0 at 1.7 GHz. A. CP Mechanism of the Proposed Dual-layer Antenna

(a)

(b)

(c)

(d)

Fig. 13. Circular polarization mechanism of the Antenna-5. (a) quadrant-2 (Region-1), (b) quadrant-3 (Region-2), (c) quadrant-4 (Region-3), and (d) quadrant-1 (Region-4). 0

14

Axial Ratio (dB)

-5

S11(dB)

-10 -15 Region 1 Region 2 Region 3 Region 4

-20 -25

Region 1 (RHCP) Region 2 (LHCP) Region 3 (RHCP) Region 4 (LHCP)

12 10 8 6 4 2

-30

0 1.6 1.8 2.0 2.2 2.4 3.4 3.6 3.8 4.0

2.0

2.2

3.6

3.8

4.0

Frequency (GHz)

Frequency (GHz)

(a)

(b)

4.2

4.4

Fig. 14. Simulation results on the placement of CSRR loaded mushroom unit cells in different quadrant. (a) Input reflection coefficient (S11) and (b) ARBW responses.

Similarly to Fig. 8(a)–(d), the placement of metaresonator from Region-1 to Region-4 for different quadrants are depicted in Fig. 13(a)–(d). The corresponding input reflection coefficient and ARBW responses are plotted in Fig. 14(a)–(b) respectively. The metaresonator placed in the quadrant-2 (Region-1) and quadrant-3 (Region-2) near to the probe feed generates RHCP and LHCP radiation at the first ARBW bands at 2.05 GHz are plotted in Fig. 14(b). Similarly, Fig. 13(c) and 13(d) depicts the placement of unit cell in quadrant-4 (Region-3) and quadrant-1 (Region-4) results in RHCP and LHCP radiations at 2.06 GHz and 2.1 GHz respectively as depicted in Fig. 14(b). Also due to RIS loading, the 16

best CP band is obtained in Fig. 13(d) [see the results of Region-4 in Fig. 14(a) and 14(b)] with LHCP radiation. ωt = 0°

ωt = 90°

ωt = 180°

ωt = 270°

(a)

(b)

(c)

(d)

Y Z

X

Fig. 15. Simulated surface current distribution of the MTM CP antenna at 2.06 GHz. (a) ωt = 0°, (b) ωt = 90°, (c) ωt = 180°, and (d) ωt = 270° (The dotted lines are indicating the direction of surface current distribution)

ωt = 0°

ωt = 90°

ωt = 180°

ωt = 270°

(a)

(b)

(c)

(d)

Y Z

X

Fig. 16. Simulated surface current distribution of the MTM CP antenna at 3.78 GHz. (a) ωt = 0°, (b) ωt = 90°, (c) ωt = 180°, and (d) ωt = 270°, (The dotted lines are indicating the direction of surface current distribution)

To confirm the CP radiation mechanism, the surface currents of RIS loaded antenna at 2.10 GHz are depicted in Fig. 15(a)–(d). Surface current distribution at excitation phase ωt = 0° is due to the current flowing through the upper left side and concentrates more in the diagonal region, which is equal in magnitude and opposite right lower side diagonal direction to that of ωt = 180°. In the case of ωt = 90° current flowing through the upper right side and more concentration in the diagonal region, which is equal in magnitude and opposite lower left side to that of ωt = 270°. It is true for the case of the designed antenna and hence the criterion for CP is satisfied at 2.10 GHz. Similarly Fig. 16(a)–(d) depicts the surface current distribution of the antenna at 3.78 GHz. Surface current distributions at excitation phase ωt = 0° and ωt = 180° showing the equal magnitude of currents and move in the opposite direction. Similarly at ωt = 90° and ωt = 270°equal magnitudes of currents and opposite directions. Hence the criteria for CP is satisfied at 3.78 GHz.

17

B. Mode Analysis of the Proposed Dual-layer Antenna 1.66 GHz

3.78 GHz

(a)

(b)

2.0 GHz

2.18 GHz

Y Z

X

+ Y Z

X

(c)

(d)

Fig. 17. Mode analysis of the proposed antenna at different frequency bands. (a) 1.66 GHz, (b) 3.78 GHz, (c) 2.0 GHz, and (d) 2.18 GHz.

For the case of normal a patch antenna, the surface current distribution at the fundamental modes forms a half-wavelength resonance and hence the fundamental mode of the conventional patch oriented in the x-direction is the TM10 mode and in y-direction is TM01 mode [33]. The surface current distributions on the patch at 1.66 GHz are depicted in Fig. 17(a). It is observed that the current distributions around the patch oriented in y-direction which allows TM01 mode to be supported at 1.66 GHz. The current distribution depicted at 3.78 GHz in Fig. 17(b) specifies that there is one-half wavelength variations both in x-direction as well as y-direction reveals the antenna supports TM11 mode. From Fig. 15(a)–(d), it is observed that the CP wave is generated due to the splitting the degenerate TM modes into TM01 (ωt = 0° and ωt = 180°) and TM10 (ωt = 90° and ωt = 270°). The TM01 produces an electric far-field Ey which is linearly polarized in y-direction and TM10 produces an electric far-field Ex which is linearly polarized in x-direction. Hence it is evident from the Fig. 17(c) and 17(d) the antenna allows TM01 mode at 2.0 GHz and TM10 at 2.18 GHz due to the two orthogonally polarized modes. C. Parametric Studies of the proposed Antenna-5 To analyze the performance of the antenna on various parameters, a parametric study is conducted by varying the dimensions of the CSRR mushroom via radius (r) from 0.2 mm to 0.6 mm as depicted 18

0

-5

-5

-10

-10 S11(dB)

S11(dB)

0

-15 -20

-30 -35

S

-25

Variation of r r = 0.2 mm r = 0.3 mm

3.8

-20 small variation of R1 R1 = 2.0 mm R1 = 2.5 mm

-25

r = 0.4 mm r = 0.5 mm r = 0.6 mm

1.4 1.6 1.8 2.0 2.2 2.4 3.6

-15

-30 -35

1.6 1.8 2.0 2.2 2.4 3.4 3.6 3.8 4.0 4.2

4.0

Frequency (GHz)

Frequency (GHz)

(b)

0

0

-5

-5

-10

-10

-15

-15

S11(dB)

S11(dB)

(a)

-20 -25 -30 -40 1.4

1.6

1.8

-20 Wa = 6.0 mm Wa = 6.5 mm Wa = 7.0 mm Wa = 7.5 mm Wa = 8.0 mm

-25 -30

a = 0.4 mm a = 0.9 mm a = 1.4 mm

-35

a = 1.9 mm a = 2.4 mm

2.0

2.2 3.6

3.8

-35 -40

4.0

1.6 1.8 2.0 2.2 2.4 3.4 3.6 3.8 4.0 4.2

Frequency (GHz)

Frequency (GHz)

(c)

(d)

15

12 r = 0.2 mm r = 0.4 mm r = 0.6 mm r = 0.8 mm r = 1.0 mm

12 9

Axial Ratio (dB)

Axial Ratio (dB)

R1 = 3.0 mm R1 = 3.5 mm R1 = 4.0 mm

6 3 0

R1 = 2.0 mm R1 = 2.5 mm R1 = 3.0 mm R1 = 3.5 mm R1 = 4.0 mm

9 6 3 0

2.0

2.2

3.6

3.8

4.0

2.0

Frequency (GHz)

2.1

2.2

3.7

3.8

3.9

4.0

Frequency (GHz)

(e)

(f)

Fig. 18. Parametric studies. (a) variation of via radius r on input reflection coefficient, (b) variation of CSRR outer radius R1 on input reflection coefficient, (c) variation of a on input reflection coefficient, (d) variation of RIS square dimensions on input reflection coefficient, (e) variation of via radius r on AR, and (f) variation of CSRR outer radius R1 on AR.

in Fig. 18(a). It is identified from the S11 response, the variation of r shifts the first resonance towards higher frequency. There is no frequency shift in the the second and third resonance except some matching differences in the second band. After optimization and analysis, an optimum value of r = 0.4 mm is selected. Similarly, Fig. 18(b) depicts the variation of C-CSRR outer ring radius R1 from 2 mm to 4 mm. It is observed that the variation of R1 shifts the resonance towards lower frequency and an optimized value of R1 = 3.5 mm is obtained. There is no variation on the second and third resonance. Fig. 18(c) depicts the variation of spacing between the CSRR mushroom unit cell and square patch. 19

Lesser spacing will result in lesser coupling and more coupling observed due to more spacing in the first band. Also due to gap a, the third resonance shifts slightly and an optimum value of a = 1.4 mm is chosen. Fig. 18(d) depicts the variation of S11 characteristics on varying RIS width Wa. It is observed that increasing Wa shifts the second and third resonances towards lower frequency and optimum result obtained at Wa = 7 mm is obtained. Hence it can be concluded that the third resonance is controlled by varying the RIS unit cell dimensions. To analyze the variation of CP radiation, Fig. 18(e) depicts the variation of ARBW on via radius r and it identifies that increase in r shifts the AR band towards higher resonance and no change in the second AR band at 3.78 GHz. Similarly, Fig. 18(f) depicts the variation of R1 on AR band and it represents that increase in R1 leads to shifts the AR band towards higher frequencies. Hence, from Fig.18(e) and 18(f) verifies that the second AR band at 3.78 have no role in the C-CSRR mushroom loadind. From Fig. 18(d) it can be understood from the S11 plot, the ARBW response at 3.78 GHz is controlled by the RIS unit cell dimensions. D. Equivalent Circuit Modelling of the proposed Dual Layer Antenna The equivalent circuit of the antenna at different stages is designed based on the circuit diagram explained in ref. [34] as shown in Fig. 19(a)–19(c). The antenna is a dual-layer configuration on which the top layer is a dual-polarized antenna. The bottom layer is RIS based MS for enhancing the antenna performances. Fig. 19(a) depicts the equivalent circuit of the proposed dual-polarized antenna. Consider Fig. 7(a), the circuit diagram consists of series inductance (LP) represents the rectangular patch of dimension L1 × W1. The capacitor (CC1) denotes the gap (a) between edge of the patch and unit cell. The capacitor (Cg1 and Cg2) denotes the coupling capacitance due to the top radiating patch and ground plane. The CSRR is represented by dotted red lines. The C-CSRR is represented [26] by the parallel combination of inductors (Lcsrr1 and Lcsrr2) which are parallel with capacitor Ccsrr. The inductor (Lvia) is represented by the via of radius r = 0.4 mm connecting in the middle of C-CSRR and ground plane represented by dotted green lines. The tuned values of inductors and capacitors for Fig. 19(a) are Lp = 12.05 nH, Lcsrr1 = Lcsrr2 = 3.9 nH, Lvia = 1.6 nH, Cc1 = 0.25 pF, Cc2 = 0.578 pF, Cg1 = 0.198 pF, Cg2 = 2.357 pF, and Ccsrr = 0.523 pF. The comparison between the CST simulated and ADS simulated input reflection coefficient responses are depicted in Fig. 20(a) are in good agreement with each other. Considering Fig. 11(c), the schematic equivalent circuit of RIS based MS is depicted in Fig. 19(b), 20

where Lm is represented by the square unit cell and Cm represents the spacing between square unit cells. The Capacitor (Cmg) is due to the coupling capacitance between RIS square cells and the ground plane. Here a single square cell is represented by dotted green colour. CSRR

Ccsrr

Square patch

LP

Cc1

Cc2

Via

Lcsrr/2 Lcsrr/2

Lvia

Cg1

Cg2

(a) Single unit cell

Lm

Lm

Cm

Cmg

Lm

Cm

Cmg

Cmg

Cm

Cmg

(b) Radiator Antenna

Cg2

Cc2

FR-4 substrate RIS metasurface

Cmg H1

H2

LP

Lm

Lcsrr/2

Ccsrr

Lcsrr/2

Lvia

Cc1

FR-4 substrate

Cmg

Cmg Cm

Cg1

Lm H1

Cmg H2

(c) Fig. 19. Equivalent circuit diagram model. (a) Dual polarized antenna depicted in Fig. 6, (b) RIS based MS represented in Fig. 8, and (c) final dual-polarized dual-layer antenna represented in Fig. 1

Fig. 19(c) represents the final circuit diagram of the antenna represented in Fig. 1, by combining the radiator equivalent circuit in Fig. 19(a) and RIS-MS in Fig. 19(b). Here the blocks H1 and H2 21

represent the FR-4 substrate [33] in the top radiator and bottom MS layer. The series resistance (R) and shunt conductance (G) for representing radiation losses and conduction loss are neglected. The proposed antenna circuit diagram depicted in Fig. 19(c) is simulated using keysight ADS software. The tuned values of inductors and capacitors are Lp = 11.709 nH, Lcsrr1 = Lcsrr2 = 3.5 nH, Lvia = 1.5 nH, Lm = 3.724 nH, Cc1 = 0.4 pF, Cc2 = 0.596 pF, Cg1= 0.458 pF, Cg2 = 4.868 pF, Ccsrr = 0.86 pF, Cm = 1.59 pF, and Cmg = 1.541 pF. The comparison between the CST and ADS simulated S11 responses depicted in Fig. 20(b) are in good agreement for the three bands. The first, second and third resonance at 1.66 GHz,

0

0

-5

-5

-10

-10

-15

S11(dB)

S11(dB)

2.13 GHz and 3.79 GHz is approximately close to the ADS simulated results depicted in Table. II.

Antenna - 3

-20 -25

-15

Antenna - 4

-20 Equivalent Ckt. CST Simulated

-25

Equivalent Ckt. CST Simulated

-30

-30 1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

1.5

2.0

2.5

3.0

Frequency (GHz)

Frequency (GHz)

(a)

(b)

3.5

4.0

Fig. 20. Comparison between the CST simulated results and ADS equivalent circuit analysis. (a) input reflection coefficient of Antenna-3, and (b) input reflection coefficient of Antenna-4.

V. EXPERIMENTAL RESULTS AND DISCUSSIONS OF THE FINAL ANTENNA PROTOTYPE The final antenna prototype was fabricated using the photolithographic process and far-field measurements are done inside the anechoic chamber using N9925A vector network analyzer. The measured 10-dB impedance bandwidths of the antenna are 60 MHz (1.66–1.72 GHz), 260 MHz (1.94– 2.20 GHz), and 190 MHz (3.68–3.87 GHz) corresponding to fractional bandwidths of 3.52%, 13.0%, and 5.02% for the three bands respectively as depicted in Fig. 21(a). The equivalent circuit depicted in Fig. 19(c) was simulated in keysight ADS software to obtain the input reflection coefficient and compared with the simulated and measured results are plotted in Fig. 21(a). A good agreement can be observed with equivalent circuit simulation, CST simulation, and measurement results. The Table. II shows the comparison of CST simulation, circuit simulation and measurement results. Fig. 21(b) shows the measured 3-dB ARBW of 60 MHz (2.08–2.14 GHz) and 70 MHz (3.75–3.82 GHz) with a fractional bandwidths of 2.85%, and 1.85% for the second and third band respectively. There is a small variation of simulated and measured ARBW is observed due to the inaccuracy in fabrication and cable losses. 22

12

0

S11(dB)

-10

Axial Ratio (dB)

-5

-15 -20 -25

BW = 3.52% BW = 13.0%

BW = 5.02%

-30 1.5

2.0

2.5

3.0

BW = 3.75 - 3.82 GHz (1.85%)

BW = 2.08 - 2.14 GHz (2.85%)

Measured CST Simulated Equivalent Ckt.

3.5

4.0

9 6 Measured Simulated

3 0 2.0

2.1

2.2

3.7

3.8

3.9

Frequency (GHz)

Frequency (GHz)

(a)

(b)

Fig. 21. Measured and simulated results of the antenna (a) Input reflection coefficient (S11), and (b) axial ratio with frequency.

TABLE. II COMPARISON BETWEEN THE CST SIMULATED, EQUIVALENT CIRCUIT SIMULATED, AND MEASURED RESULTS OF THE PROPOSED ANTENNA Frequency Bands First Band Second Band Third Band

CST simulated 1.66 2.05 3.78

ADS circuit simulation 1.66 2.13 3.79

Measurement 1.70 2.10 3.78

The far-field radiation characteristics of the proposed antenna are studied inside the anechoic chamber. Fig. 22(a)–22(c) depicts the measured and simulated two-dimensional radiation patterns plotted at the two principal planes (yz–plane and xz–plane) at the center frequencies of 1.7 GHz, 2.1 GHz, and 3.78 GHz respectively. As can be observed from the pattern in Fig. 22(a) at 1.7 GHz, the antenna shows linear polarization with directional patterns in both the planes and back lobe is reduced due to the RIS ground plane and the co-polarized fields in boresight direction are stronger than crosspolarized fields. Fig. 20(b) at 2.1 GHz, the antenna shows directional pattern with left-handed circular polarization (LHCP) radiation is observed at both the planes in +Z direction. It is to be observed that at 2.1 GHz the antenna provides wider beamwidth CP radiation (ARBW < 3 dB) for theta = ±580 and for all phi values as depicted in Fig 23(a). Also positive gain values are obtained for theta = ±600 and for all phi values at 2.1 GHz is plotted in Fig 23(b). Hence the antenna is suitable for working in small satellite application systems. Fig. 22(c) depicts the radiation pattern at 3.78 GHz, which shows righthanded circular polarization radiation. The antennas shows monopole like radiation pattern with a null in broadside direction of the pattern for yz-plane and xz-plane and maximum gain is around θ = 3100 is observed. This can be observed in the mode analysis depicted in Fig. 17(b), the currents are concentrated on the edges and no currents in the middle part. Hence the maximum radiation is obtained at the sides and null is observed in the broadside direction. 23

yz - plane 330

0

xz - plane

0 330

30

-10

-20

-25

270

90

240

210

0

330

Theta = +58 Theta = -58

-10

300

(a)

30

-5

60

270

120

210

Theta = +58 60

-25

0

(b)

30

90

240 LHCP Measured LHCP Simulated RHCP Measured RHCP Simulated

150 180

120

210

150 180

330

0

0 30

-5

-5 -10

-10

300

60

60

-15

-15

-20

-20 270

30

-20 90

240

300

0

-15

-25

0

0 -10

300

-20

330

150 180

-5

-15 270

90

120

Co-pol Measured 210 Co-pol Simulated Cross-pol Simulated Cross-pol Measured

150 180

Theta = -58

-25

240

120

0

60

-15

-20

330

30

-10

300

60

-15 270

0

-5

-5 300

0

-25

270

90

240

150 180

90

240

120

210

-25

RHCP Measured RHCP Simulated LHCP Measured LHCP Simulated

120

210

150 180

(c)

Fig. 22. Measured and simulated radiation patterns (a) 1.7 GHz, (b) 2.1 GHz, and (c) 3.78 GHz.

12

Phi = 00 Phi = 900

9

Phi = 1800 0 Phi = 270

6

4

Gain (dBi)

Axial Ratio (dB)

15

ARBW < 3 dB for Theta = ± 580 for all Phi values

3

2 0 -2 -4

0 -120 -90 -60 -30

0

30

60

90 120

Theta (Degrees)

0 Phi = 0 0 Phi = 90 0 Phi = 180 0 Phi = 270 Gain Positive values for Theta = ± 600 for all Phi values

-6 -120 -90 -60 -30

0

30

60

90 120

Theta (Degrees)

(a)

(b) 0

Fig. 23. Simulated results of the final antenna at 2.1 GHz (a) Axial ratio (theta = ± 58 and for all phi = 00 to 2700), and (b) gain (theta = ± 580 and for all phi = 00 to 2700).

24

Radiation Efficiency (%)

6 Gain (dBi)

100

Without RIS Simulated Without RIS Measured With RIS Simulated With RIS Measured

8

4 2 0 -2

80 60 40 20 0

Without RIS simulated With RIS simulated With RIS measured

-20

1.6 1.7 1.8 1.9 2.0 2.1 2.2

3.7 3.8 3.9

Frequency (GHz)

1.6 1.7 1.8 1.9 2.0 2.1 2.2

3.7 3.8 3.9

Frequency (GHz)

(a)

(b)

Fig. 24. Simulated and measured results with and without RIS (a) gain, and (b) radiation efficiency.

Fig. 24(a) shows the measured and simulated gain plots of the antenna with and without RISMS measured by gain comparison method. For antenna without RIS ground plane, the antenna shows measured gain values of -1.08 dBi, 1.15 dBic, and -1.3 dBic at the three operating frequency of 1.7 GHz, 2.1 GHz, and 3.78 GHz respectively. When RIS is incorporated with antenna, the gain enhanced to 1.20 dBi, 4.72 dBic and 3.48 dBic for the three operating bands. The radiation efficiency of the antenna is measured using wheeler cap method [35] is depicted in Fig. 24(b). For antenna without RIS, lesser efficiency of 2.52%, 36%, and 23% are obtained for the three consecutive bands. When the RIS is incorporated, the simulated radiation efficiency of 45%, 70%, and 69% are obtained for the three bands and measured radiation efficiency of 45.3%, 69.2% and 68.77% are obtained at 1.66 GHz, 2.0 GHz and 3.66 GHz for the three bands respectively. TABLE. III COMPARISON BETWEEN THE MEASURED RESULTS OF THE ANTENNA WITH AND WITHOUT RIS METASURFACE Antenna Parameters Antenna Size (mm3) Frequency (GHz) ka (radiating element) Imp. Bandwidth (%) ARBW (%) Gain (dBi) Radiation efficiency Radiation Pattern Polarization

Radiator Antenna (Ant. 3) 48 × 48 × 1.6 2.31 4.46 0.99 * 6.14 2.35* 1.17*



0.9 36* Unidirectional Circular (LHCP)

-1.5 23* Bidirectional Linear

Radiator Antenna with RIS-MS (Ant. 4) 48 × 48 × 4.8 1.70 2.10 3.78 0.72 3.52 13.0 5.02 2.85 (θ = ±580, – 1.85 all ϕ) 1.20 4.72 3.48 45.3 69.2 68.77 Unidirectional Unidirectional Bidirectional Circular Circular Linear (LHCP) (RHCP)

*

indicated simulated results

Table. III shows the comparison of the antenna with and without RIS. It is observed that the RIS loading can generate multi-band characteristics with improved bandwidth, enhanced gain, ARBW and 25

acceptable radiation efficiency with antenna compactness. Table. IV shows the comparative study of existing MTM dual-band CP antennas, existing dual-polarized multiband antennas based on size, impedance bandwidth, ARBW, polarization, gain, and radiation efficiency with the measured performances of the proposed antenna. TABLE. IV COMPARISON BETWEEN THE PROPOSED ANTENNA AND EXISTING DUAL BAND CP ANTENNAS, AND MULTIBAND DUAL POLARIZED METASURFACE BASED ANTENNAS Ref. Number

Configuration Used

No: of Bands

[9]

CRLH mushroom structure and RIS

3

Freq. (GHz) 2.42 3.84 3.37 1.95 2.61 2.30 2.56 2.89 3.82 2.86 3.11 1.38 1.57 2.56 5.37 1.33 1.88 2.41 1.225 1.575 1.88

Antenna physical size (mm3) 34 × 34 × 3

Imp. BW (%) 1.61 3.27 3.08 1.28 5.3 4.56 2.15 2.975 0.62 1.75 2.57 2 1 15.6 9.3 1.88# 3.24# 10.03# – – 7.5

AR BW (%) – – – – 0.7 1.2 – 0.76 0.18 1.05 1.61 – 1.27 – – – – 1 >2 >2 2.7

Gain (dBi) 0.27 3.31 4.45 -6.9 -1.1 2.0 2.27 6.26 6.97 4.15 4.77 2 7* 7.2* 7.3* 2.1 0.6 5.7 6* 7* 3.95

Radiation Efficiency (%) 43.7 69.8 75.5 28 58 NA NA 82 65 67.6 69.6 NA NA 65 65 27 45 66 NA NA NA

CRLH-TL with 22 × 24.8 × 2 square CSRR 1.6 Patch with square 60 × 60 × [12] 2 CSRR mushroom 1.6 Trimmed patch 60 × 60 × [13] 2 with mushroom 3.175 Metaresonator 43.5 × 43.5 [15] 2 and RIS ×2 Dual mode patch 125 × 125 [16] 2 with AMC × 6.50 Dual polarized 104 × 104 [17] 2 patch with AMC × 11 Patch with 70 × 70 × mushroom loaded [18] 3 2.5 with RIS Stacked concentric 140 × 140 [19] 2 annular ring patch × 12 Monopole with 52 × 52 × [20] 2 2×2 CSRR unit 6.94 2.50 12.6 2 5.29 NA cells Annular slot 1.23 3.2 1.62 8* 97.5 150 ×150 × [21] radiator with HIS 2 18 1.58 3.1 3.16 7.5* 85.7 ground plane – 1.70 3.52 1.20 45.3 MTM inspired Prop. 48 × 48 × CP antenna 3 2.10 13.0 2.85 4.72 70.2 Antenna 4.8 loaded with RIS 3.78 5.02 1.85 3.48 69.7 Note: The electrical size of antenna configurations explained in the above table are calculated on the corresponding wavelength at first resonating frequency. # Bandwidth is calculated at -6 dB points of S11 * Higher value of antenna gain is obtained due to antenna larger profile and higher size [11]

Polarization Linear Linear Linear Linear RHCP LHCP Linear LHCP LHCP LHCP RHCP Linear RHCP Linear Linear Linear Linear NA RHCP RHCP RHCP LHCP RHCP RHCP Linear LHCP RHCP basis of

A. Chu-Limit and Harrington Bound for Electrically Small Antennas For electrically compact antennas, Wheeler [36] firstly explained that the antenna dimensions should be lower than radian the sphere (λ⁄2𝜋). Later Chu [37] introduces the Chu limit, which describes a lower limit on quality factor (𝑄) depends upon the antenna’s physical size which reduces its full 26

attainable bandwidth (𝐹𝐵𝑊𝑚𝑎𝑥 ). Later Mc Lean [38] modified this work and conferred the newly revised form for 𝐹𝐵𝑊𝑚𝑎𝑥 and 𝑄 explained in eqn. (7) and eqn. (8) 𝑄𝑚𝑖𝑛 = 𝐹𝐵𝑊𝑚𝑎𝑥 =

1 𝑘 3 𝑎3

+

1 𝑘𝑎

𝑉𝑆𝑊𝑅−1 𝑄𝑚𝑖𝑛 √𝑉𝑆𝑊𝑅

(7) (8)

Electrically compact antennas are extremely small which faces the problem of narrow impedance bandwidth, which identifies that it suffers from impedance matching problems. Also when the top radiating element and bottom ground plane are very near, the alternating current moving on the surface of antenna obtains resistance from the induced current on the ground plane. Due to this, higher energy in the nearfield tends to lesser gain and poor radiation efficiency. Therefore, the maximum realizable gain of an electrically compact antenna can be obtained by Harrington bound [39] represented in eqn. (9) 𝐺𝑑𝐵𝑖 = 10𝑙𝑜𝑔10 ((𝑘𝑎)2 + 2𝑘𝑎)

(9)

For the proposed MTM inspired antenna depicted in Fig. 1, the first resonating frequency at 1.66 GHz. The calculated value of 𝑘 = 35.58 rad/m, 𝑎 = 20.5 mm and hence 𝑘𝑎 = 0.72 < 1, the proposed antenna is electrically small. Using eqn. (7) and eqn. (8) for 𝑉𝑆𝑊𝑅 = 2, 𝑄𝑚𝑖𝑛 = 4.06, the maximum obtainable bandwidth 𝐹𝐵𝑊𝑚𝑎𝑥 = 17.41% and measured fractional bandwidth is 3.52%. Also, a maximum feasible gain of 2.91 dBi is obtained using eqn. (9) and measured gain of 1.20 dBi is obtained.

VI. CONCLUSION A compact MTM inspired multiband antenna integrated with RIS based MS for gain enhancement and dual-band CP radiation is investigated. Antenna compactness and wider AR beamwidth are achieved by the combination of C-CSRR mushroom-based metaresonator and RIS. The antenna obtains a miniaturized radiating element size of 0.16 λ0 × 0.16 λ0 at 1.7 GHz due to the negative permeability property of metaresonator with an electrically compact size of 𝑘𝑎 = 0.72. An area reduction of approximately 52.32% is obtained when compared to the conventional square patch antenna. Due to the RIS loading, the overall antenna gain increased to 1.20 dBi, 4.72 dBic, and 3.48 dBic for the three operating bands. The contribution of the design is, it can generate multi-band CP radiation with wider AR beamwidth, improved gain, improved impedance bandwidth, and acceptable radiation efficiency by keeping the antenna size smaller compared with similar types of existing antenna designs. The antenna 27

provides a wider AR beamwidth for CP radiation (ARBW<3) with positive gain values for theta = ±580 and for all phi values at 2.05 GHz. The antenna shows multifunctional radiation performances such as the first band provide directional pattern with LP radiation, the second band provides directional pattern with LHCP radiation and the third band provides a bidirectional radiation with RHCP radiation performance. Hence, the antenna is suitable for working in 1.7 GHz LTE (1.71–1.88 GHz), 2.2 GHz UMTS (1.88–2.2 GHz), 3.6/3.8 GHz WiMAX (3.4–3.8 GHz) and small satellite application systems at 2.1 GHz.

ACKNOWLEDGMENT This research work is supported by Science and Engineering Research Board (SERB), Department of Science and Technology (DST), Government of India under grant number EEQ/2016/000023.

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30

List of Figures Title of the paper: Dual-layer and Dual-polarized Metamaterial Inspired Antenna using CircularComplementary Split Ring Resonator Mushroom and Metasurface for Wireless Applications

C-CSRR loaded Mushroom Coaxial feeding Patch antenna with C-CSRR loaded mushroom

Z H1 H2 Y

Top Radiator Layer Bottom RIS Layer

X

Z

SMA connector Via

X

Y

RIS unit cells

(a)

(b)

Fig. 1. Triple-band dual-layer and dual-polarized MTM inspired antenna. (a) Two-dimensional front view, and (b) Threedimensional top view.

Square Patch

Square Patch with slot

C-CSRR loaded Modified Mushroom

RIS Loaded Antenna

Feed point (Xf,Yf)

L1

a W1 Antenna 1

(a)

Antenna 2

Antenna 3

Antenna 4

(b)

(c)

(d)

Fig. 2. Design stages of the proposed MTM inspired antenna. (a) microstrip square patch antenna, (b) square patch antenna with a square slot, (c) C-CSRR mushroom unit cell loaded antenna, and (d) MTM inspired antenna loaded with RIS.

31

0 -5

Axial Ratio (dB)

12

S11(dB)

-10 -15 -20 Antenna 1 Antenna 2 Antenna 3 Antenna 4

-25 -30 -35 1.5

2.0

2.5

3.0

3.5

4.0

Antenna 2 Antenna 3 Antenna 4

9 6 3 0

4.5

5.0

2.0

Frequency (GHz)

2.1

2.2

2.3

3.7

3.8

3.9

4.0

Frequency (GHz)

(a)

(b)

Fig. 3. proposed antenna design stage responses from Antenna-1 to Antenna-4 (a) input reflection coefficient (S11), and (b) ARBW.

W4

Conventional square unit cell L4

Conventional mushroom Via r = 0.3 mm

R2

R1

G3

(Xl, Yl) G2

Top view

G1

(a)

H1

Via

Side view (b)

(c)

Fig. 4. Design of C-CSRR loaded mushroom unit cell (a) simple square patch, (b) conventional mushroom unit cell, and (c) MTM inspired C-CSRR loaded mushroom unit cell.

32

Port-2

CSRR

Transmission Line

Microstrip feed via

Transmission Line

Ccsrr

L1

Lcsrr/2

L2

Via

Lcsrr/2 Lvia

C1

C2

Port-1

CSRR Mushroom unit cell

-2 Port

-1 Port Z

Y X

0 -5 -10 -15

First order

-20 Second order Simulated S11 Simulated S12 Equivalent ckt. S11 Equivalent ckt S12

-25 -30 -35 -40

Ground plane

1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5

FR-4 substrate (b)

Frequency (GHz)

(c)

Conventional mushroom unit cell

4.5 4.0

Conventional case CSRR Mushroom

3.5

RH Region

3.0 CSRR Mushroom unit cell

Permittivity (ε )

Frequency (GHz)

5.0

2.5 LH Region

2.0 0

20 40 60 80 100 120 140 160 180 d (degrees)

(d)

150 120 90 60 30 Real (ε) 0 -30 -60 -90 -120 1.5 1.8

Real (ε)

5 4 3 2 1 0 -1 -2 -3 -4

Real (μ)

Real (ε) Imag (ε) Real (μ) Imag (μ)

2.1 2.4 2.7 3.0 Frequency (GHz)

3.3

Permeability (μ)

Microstrip feed

Scattering Parameters (dB)

(a) via CSRR

3.6

(e)

Fig. 5. C-CSRR loaded mushroom characteristics. (a) CSRR loaded mushroom with microstrip line feed and corresponding equivalent circuit diagram, (b) Two-port set up for the analysis of CSRR mushroom, (c) Transmission and reflection coefficients, (d) dispersion diagram plots of the proposed and conventional unit cell, and (e) real and imaginary permittivity response.

33

4.3

4.1 4.0

Frequency (GHz)

Frequency (GHz)

4.2

Second mode First mode

2.3 2.2 2.1 2.0

4.2 4.1 2.4

Second mode First mode

2.3 2.2 2.1

1.9 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1

2.0 2.2 2.4 2.6 2.8 3.0 3.2 3.4 3.6 3.8

Variation of r (mm)

Variation of R1 (mm)

(a)

(b)

Fig. 6. C-CSRR mushroom unit cell characteristics. (a) variation of via radius (r), and (b) variation of CSRR outer ring radius (R1).

Via

W4

a L2 L

L1

W2 L3 Probe feed

W Z

X

R1

L4

(Xf,Yf)

W1

Y

R2

(Xl, Yl) G2 Patch

G1 C-CSRR Copper Substrate

(a)

(b)

Fig. 7. Multiband MTM inspired antenna (a) schematic view of the multiband MTM inspired antenna with an enlarged view of C-CSRR mushroom unit cell, and (b) fabricated antenna prototype.

34

Region-1

Square Patch

Region-4 Metaresonator

Feed point

Region-3

Region-2 (a)

(b)

(c)

(d)

Fig. 8. Studies on the placement of C-CSRR mushroom unit cell in different quadrant. (a) quadrant-2 (Region-1), (b) quadrant-3 (Region-2), (c) quadrant-4 (Region-3), and (d) quadrant-1 (Region-4).

0

14 Axial Ratio (dB)

-5 S11(dB)

-10 -15 -20

Region 1 Region 2 Region 3 Region 4

-25 -30 2.4

2.6 4.0

4.2

4.4

12 10 8 6 4 2

-35 2.0 2.2

Region 1 (RHCP) Region 2 (LHCP) Region 3 (RHCP) Region 4 (LHCP)

4.6

4.8

0 2.10

2.15

2.20

2.25

2.30

2.35

Frequency (GHz)

Frequency (GHz)

(a)

(b)

Fig. 9. Simulation results on the placement of CSRR loaded mushroom unit cells in different quadrant. (a) Input reflection coefficient (S11) and (b) ARBW responses.

35

ωt = 0°

ωt = 90°

ωt = 180°

ωt = 270°

(a)

(b)

(c)

(d)

Y Z

X

Fig. 10. Simulated surface current distribution of the MTM CP antenna at 2.25 GHz (a) ωt = 0°, (b) ωt = 90°, (c) ωt = 180°, and (d) ωt = 270° (ωt represents the excitation phase).

Reflection Phase (Degrees)

Waveport Simplified eq. ckt

PMC

PEC PMC

PEC Open

Z

RIS unit cell Y X

200 1800 100 00

0 -100

0.66 - 2.77 GHz RIS operating range

-200 1

Waveport

2

3

4

5

6

Frequency (GHz)

(a)

(b)

Lb Wa G

Wb

Via

Hole for feed line

La

Y

Z

X

(c)

(d)

Fig. 11. RIS design, (a) Metasurface unit cell enclosed by the boundary conditions (floquet port), (b) reflection phase response, (c) schematic of the metasurface, and (d) fabricated MS ground plane.

36

Cm Lm Cm Lm

Cmg

Lm

Z1 Cmg

Z2

Cmg

Fig. 12. Equivalent circuit representation of the RIS based MS unit cell depicted in Fig. 11(a).

(a)

(b)

(c)

(d)

Fig. 13. Circular polarization mechanism of the Antenna-5. (a) quadrant-2 (Region-1), (b) quadrant-3 (Region-2), (c) quadrant-4 (Region-3), and (d) quadrant-1 (Region-4).

0

14

Axial Ratio (dB)

-5

S11(dB)

-10 -15 Region 1 Region 2 Region 3 Region 4

-20 -25

Region 1 (RHCP) Region 2 (LHCP) Region 3 (RHCP) Region 4 (LHCP)

12 10 8 6 4 2

-30

0 1.6 1.8 2.0 2.2 2.4 3.4 3.6 3.8 4.0

2.0

2.2

3.6

3.8

4.0

Frequency (GHz)

Frequency (GHz)

(a)

(b)

4.2

4.4

Fig. 14. Simulation results on the placement of CSRR loaded mushroom unit cells in different quadrant. (a) Input reflection coefficient (S11) and (b) ARBW responses.

37

ωt = 0°

ωt = 90°

ωt = 180°

ωt = 270°

(a)

(b)

(c)

(d)

Y Z

X

Fig. 15. Simulated surface current distribution of the MTM CP antenna at 2.06 GHz. (a) ωt = 0°, (b) ωt = 90°, (c) ωt = 180°, and (d) ωt = 270° (The dotted lines are indicating the direction of surface current distribution)

ωt = 0°

ωt = 90°

ωt = 180°

ωt = 270°

(a)

(b)

(c)

(d)

Y Z

X

Fig. 16. Simulated surface current distribution of the MTM CP antenna at 3.78 GHz. (a) ωt = 0°, (b) ωt = 90°, (c) ωt = 180°, and (d) ωt = 270°, (The dotted lines are indicating the direction of surface current distribution)

1.66 GHz

3.78 GHz

(a)

(b)

2.0 GHz

2.18 GHz

Y Z

X

+ Y Z

X

(c)

(d)

Fig. 17. Mode analysis of the proposed antenna at different frequency bands. (a) 1.66 GHz, (b) 3.78 GHz, (c) 2.0 GHz, and (d) 2.18 GHz.

38

0

-5

-5

-10

-10 S11(dB)

S11(dB)

0

-15 -20

-30 -35

S

-25

Variation of r r = 0.2 mm r = 0.3 mm

3.8

-20 small variation of R1 R1 = 2.0 mm R1 = 2.5 mm

-25

r = 0.4 mm r = 0.5 mm r = 0.6 mm

1.4 1.6 1.8 2.0 2.2 2.4 3.6

-15

-30 -35

1.6 1.8 2.0 2.2 2.4 3.4 3.6 3.8 4.0 4.2

4.0

Frequency (GHz)

Frequency (GHz)

(a)

(b) 0

-5

-5

-10

-10

-15

-15

S11(dB)

S11(dB)

0

-20 -25 -30 -40 1.4

1.6

1.8

-20 Wa = 6.0 mm Wa = 6.5 mm Wa = 7.0 mm Wa = 7.5 mm Wa = 8.0 mm

-25 -30

a = 0.4 mm a = 0.9 mm a = 1.4 mm

-35

a = 1.9 mm a = 2.4 mm

2.0

2.2 3.6

3.8

-35 -40

4.0

1.6 1.8 2.0 2.2 2.4 3.4 3.6 3.8 4.0 4.2

Frequency (GHz)

Frequency (GHz)

(c)

(d)

15

12 r = 0.2 mm r = 0.4 mm r = 0.6 mm r = 0.8 mm r = 1.0 mm

12 9

Axial Ratio (dB)

Axial Ratio (dB)

R1 = 3.0 mm R1 = 3.5 mm R1 = 4.0 mm

6 3 0

R1 = 2.0 mm R1 = 2.5 mm R1 = 3.0 mm R1 = 3.5 mm R1 = 4.0 mm

9 6 3 0

2.0

2.2

3.6

3.8

4.0

2.0

Frequency (GHz)

2.1

2.2

3.7

3.8

3.9

4.0

Frequency (GHz)

(e)

(f)

Fig. 18. Parametric studies. (a) variation of via radius r on input reflection coefficient, (b) variation of CSRR outer radius R1 on input reflection coefficient, (c) variation of a on input reflection coefficient, (d) variation of RIS square dimensions on input reflection coefficient, (e) variation of via radius r on AR, and (f) variation of CSRR outer radius R1 on AR.

39

CSRR

Ccsrr

Square patch

LP

Cc1

Cc2

Via

Lcsrr/2 Lcsrr/2

Lvia

Cg1

Cg2

(a) Single unit cell

Lm

Lm

Cm

Cmg

Lm

Cm

Cmg

Cmg

Cm

Cmg

(b) Radiator Antenna

Cg2

Cc2

FR-4 substrate RIS metasurface

Cmg H1

H2

LP

Lm

Lcsrr/2

Ccsrr

Lcsrr/2

Lvia

Cc1

FR-4 substrate

Cmg

Cmg Cm

Cg1

Lm H1

Cmg H2

(c) Fig. 19. Equivalent circuit diagram model. (a) Dual polarized antenna depicted in Fig. 6, (b) RIS based MS represented in Fig. 8, and (c) final dual-polarized dual-layer antenna represented in Fig. 1

40

0

-5

-5

-10

-10

-15

S11(dB)

S11(dB)

0

Antenna - 3

-20 -25

-15

Antenna - 4

-20 Equivalent Ckt. CST Simulated

-25

Equivalent Ckt. CST Simulated

-30

-30 1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

1.5

2.0

2.5

Frequency (GHz)

3.0

3.5

4.0

Frequency (GHz)

(a)

(b)

Fig. 20. Comparison between the CST simulated results and ADS equivalent circuit analysis. (a) input reflection coefficient of Antenna-3, and (b) input reflection coefficient of Antenna-4.

12

0

S11(dB)

-10

Axial Ratio (dB)

-5

-15 -20 -25

BW = 3.52% BW = 13.0%

BW = 5.02%

-30 1.5

2.0

2.5

3.0

3.5

BW = 3.75 - 3.82 GHz (1.85%)

BW = 2.08 - 2.14 GHz (2.85%)

Measured CST Simulated Equivalent Ckt.

4.0

9 6 Measured Simulated

3 0 2.0

2.1

2.2

3.7

3.8

3.9

Frequency (GHz)

Frequency (GHz)

(a)

(b)

Fig. 21. Measured and simulated results of the antenna (a) Input reflection coefficient (S11), and (b) axial ratio with frequency.

41

yz - plane 330

0

xz - plane

0 330

30

-10

-20

-25

270

90

240

210

0

330

Theta = +58 Theta = -58

-10

300

(a)

30

-5

60

300

270

120

210

Theta = +58 60

0

(b)

30

-25

120

210

330

0

0 30

-5

-10

-10

300

60

60

-15

-15

-20

-20 -25

270

90

240

150

-25

90

240

120

180

150 180

-5

210

90

240 LHCP Measured LHCP Simulated RHCP Measured RHCP Simulated

150 180

270

30

-20 90

240

300

0

-15

-25

0

0 -10

-20

330

150 180

-5

-15 270

90

120

Co-pol Measured 210 Co-pol Simulated Cross-pol Simulated Cross-pol Measured

150 180

Theta = -58

-25

240

120

0

60

-15

-20

330

30

-10

300

60

-15 270

0

-5

-5 300

0

RHCP Measured RHCP Simulated LHCP Measured LHCP Simulated

120

210

150 180

(c)

Fig. 22. Measured and simulated radiation patterns (a) 1.7 GHz, (b) 2.1 GHz, and (c) 3.78 GHz.

42

12

Phi = 00 Phi = 900

9

Phi = 1800 0 Phi = 270

6

4 Gain (dBi)

Axial Ratio (dB)

15

ARBW < 3 dB for Theta = ± 580 for all Phi values

3

2 0 -2 -4

0 -120 -90 -60 -30

0

30

60

0 Phi = 0 0 Phi = 90 0 Phi = 180 0 Phi = 270 Gain Positive values for Theta = ± 600 for all Phi values

-6 -120 -90 -60 -30

90 120

Theta (Degrees)

0

30

60

90 120

Theta (Degrees)

(a)

(b) 0

Fig. 23. Simulated results of the final antenna at 2.1 GHz (a) Axial ratio (theta = ± 58 and for all phi = 00 to 2700), and (b) gain (theta = ± 580 and for all phi = 00 to 2700).

Gain (dBi)

6

100

Without RIS Simulated Without RIS Measured With RIS Simulated With RIS Measured

Radiation Efficiency (%)

8

4 2 0 -2

80 60 40 20 0

Without RIS simulated With RIS simulated With RIS measured

-20

1.6 1.7 1.8 1.9 2.0 2.1 2.2

3.7 3.8 3.9

Frequency (GHz)

1.6 1.7 1.8 1.9 2.0 2.1 2.2 Frequency (GHz)

(a)

(b)

Fig. 24. Simulated and measured results with and without RIS (a) gain, and (b) radiation efficiency.

43

3.7 3.8 3.9

List of Tables Title of the paper: Dual-layer and Dual-polarized Metamaterial Inspired Antenna using CircularComplementary Split Ring Resonator Mushroom and Metasurface for Wireless Applications

TABLE I. COMPARISON OF THE SIMULATED ANTENNA PERFORMANCE FROM ANTENNA-1 TO ANTENNA-4 Radiating No: of Freq. Ka* Imp. ARBW Antenna element size Reason for the Bands Bands (GHz) value BW (%) (%) (λ03) 2.41 – – Due to square patch 0.23 × 0.23 × Antenna-1 2 1.03 0.012 4.81 6.97 – Higher-order mode 2.20 3.09 – Due to square patch 0.21 × 0.21 × Antenna-2 3 2.39 0.94 2.92 – Rectangular slot loading 0.011 4.48 2.67 – Due to Higher-order mode Merging of resonances due 2.31 6.14 1.35 to metaresonator and 0.22 × 0.22 × Antenna-3 2 0.99 slotted patch 0.011 4.46 2.35 – Due to Higher-order mode 1.66 4.27 – Due to metaresonator Merging of resonances due 2.05 0.16 × 0.16 × 14.63 2.36 to square patch and Antenna-4 3 0.71 0.02 rectangular slot Higher-order modes and 3.78 3.96 2.09 RIS metasurface *

Note: In all the antenna design stages, the ka value is calculated on the basis of the first resonating frequency. Also in Antenna-4, the final resonance at 4.55 GHz is not considered.

TABLE. II COMPARISON BETWEEN THE CST SIMULATED, EQUIVALENT CIRCUIT SIMULATED, AND MEASURED RESULTS OF THE PROPOSED ANTENNA Frequency Bands First Band Second Band Third Band

CST simulated 1.66 2.05 3.78

ADS circuit simulation 1.66 2.13 3.79

44

Measurement 1.70 2.10 3.78

TABLE. III COMPARISON BETWEEN THE MEASURED RESULTS OF THE ANTENNA WITH AND WITHOUT RIS METASURFACE Antenna Parameters Antenna Size (mm3) Frequency (GHz) ka (radiating element) Imp. Bandwidth (%)

Radiator Antenna (Ant. 3) 48 × 48 × 1.6 2.31 4.46 0.99 6.14* 2.35* 1.17*



0.9 36* Unidirectional Circular (LHCP)

-1.5 23* Bidirectional

ARBW (%) Gain (dBi) Radiation efficiency Radiation Pattern Polarization

Linear

Radiator Antenna with RIS-MS (Ant. 4) 48 × 48 × 4.8 1.70 2.10 3.78 0.72 3.52 13.0 5.02 0 2.85 (θ = ±58 , – 1.85 all ϕ) 1.20 4.72 3.48 45.3 69.2 68.77 Unidirectional Unidirectional Bidirectional Circular Circular Linear (LHCP) (RHCP)

*

indicated simulated results

TABLE. IV COMPARISON BETWEEN THE PROPOSED ANTENNA AND EXISTING DUAL BAND CP ANTENNAS, AND MULTIBAND DUAL POLARIZED METASURFACE BASED ANTENNAS Ref. Number

Configuration Used

No: of Bands

[9]

CRLH mushroom structure and RIS

3

[11] [12] [13] [15] [16] [17] [18] [19] [20]

[21] Prop. Antenna

CRLH-TL with square CSRR Patch with square CSRR mushroom Trimmed patch with mushroom Metaresonator and RIS Dual mode patch with AMC Dual polarized patch with AMC Patch with mushroom loaded with RIS Stacked concentric annular ring patch Monopole with 2×2 CSRR unit cells Annular slot radiator with HIS ground plane

2 2 2 2 2 2 3 2 2

Freq. (GHz) 2.42 3.84 3.37 1.95 2.61 2.30 2.56 2.89 3.82 2.86 3.11 1.38 1.57 2.56 5.37 1.33 1.88 2.41 1.225 1.575 1.88 2.50 1.23

2 3

Antenna physical size (mm3) 34 × 34 × 3 22 × 24.8 × 1.6 60 × 60 × 1.6 60 × 60 × 3.175 43.5 × 43.5 ×2 125 × 125 × 6.50 104 × 104 × 11 70 × 70 × 2.5 140 × 140 × 12 52 × 52 × 6.94

1.58

150 ×150 × 18

1.70 2.10

48 × 48 × 4.8

45

Imp. BW (%) 1.61 3.27 3.08 1.28 5.3 4.56 2.15 2.975 0.62 1.75 2.57 2 1 15.6 9.3 1.88# 3.24# 10.03# – – 7.5

AR BW (%) – – – – 0.7 1.2 – 0.76 0.18 1.05 1.61 – 1.27 – – – – 1 >2 >2 2.7

0.27 3.31 4.45 -6.9 -1.1 2.0 2.27 6.26 6.97 4.15 4.77 2 7* 7.2* 7.3* 2.1 0.6 5.7 6* 7* 3.95

Radiation Efficiency (%) 43.7 69.8 75.5 28 58 NA NA 82 65 67.6 69.6 NA NA 65 65 27 45 66 NA NA NA

12.6

2

5.29

NA

LHCP

3.2

1.62

8*

97.5

RHCP

3.1

3.16

7.5*

85.7

RHCP

3.52 13.0

– 2.85

1.20 4.72

45.3 70.2

Linear LHCP

Gain (dBi)

Polarization Linear Linear Linear Linear RHCP LHCP Linear LHCP LHCP LHCP RHCP Linear RHCP Linear Linear Linear Linear NA RHCP RHCP RHCP

MTM inspired 1.85 69.7 RHCP CP antenna 3.78 5.02 3.48 loaded with RIS Note: The electrical size of antenna configurations explained in the above table are calculated on the basis of corresponding wavelength at first resonating frequency. # Bandwidth is calculated at -6 dB points of S11 * Higher value of antenna gain is obtained due to antenna larger profile and higher size

46

Title: Dual-layer and Dual-polarized Metamaterial Inspired Antenna using CircularComplementary Split Ring Resonator Mushroom and Metasurface for Wireless Applications

Biographies: Mohammad Ameen was born in Kerala, India on October 07, 1990. He received B.Tech and M.Tech degree in Electronics and Communication Engineering from Mahatma Gandhi University, Kottayam, Kerala in 2012 and 2014 respectively. He has an industrial experience of working as a Graduate Trainee at Centre for Development of Advanced Computing (C-DAC), Government of India, Kerala. He worked as a Senior Research Fellow in Council of Scientific and Industrial Research (CSIR), Government of India funded project at Centre for Research in Electromagnetics and Antennas, Cochin University of Science and Technology, Kerala during 2015-2017. Currently, he is a Senior Research Fellow in DST(SERB) sponsored project, Government of India and pursuing Ph.D. from Department of Electronics Engineering, Indian Institute of Technology (Indian School of Mines). Dhanbad, Jharkhand, India. He is a potential reviewer of many journals and conferences such as IEEE Access, Microwave and Optical Technology Letters, International Journal of RF & Microwave Computer Aided Engineering etc. His current research interests involve Metamaterials, Electromagnetic Bandgap Structures, Artificial Magnetic Conductors, Frequency Selective Surfaces, MIMO antennas, Electrically small antennas, Flexible antennas, Metamaterial based sensors and its applications.

Dr. Raghvendra Kumar Chaudhary is working as an Assistant Professor at Department of Electronics Engineering, Indian Institute of Technology (ISM), Dhanbad, India. He did his Ph.D. from IIT Kanpur, India in Jan. 2014, the M.Tech. degree from IIT(BHU) Varanasi, India, in 2009 and the B.Tech. degree from UIET Kanpur India, in 2007. His current research interests include Dielectric Resonator Antenna, Metamaterial Antenna, MIMO Antenna, and Metamaterial Absorber. Dr. Chaudhary has authored over 100 international Journal papers and 80 conference papers. Dr. Chaudhary has served as Chair, IEEE Student Branch, Uttar Pradesh Section, in 2012 and 2013, respectively. He was recipient many best paper awards in various conferences such as IEEE APACE Malaysia, PIERS Singapore, PIERS Italy, ATMS, India, etc. He has also started 2-Year M.Tech. Course Program in RF & Microwave Engineering at Department of Electronics Engineering, IIT(ISM) Dhanbad. He is Associate Editor of IET Microwave Antennas and Propagation, Microwave and Optical Technology Letters, IEEE Access, Senior Member of IEEE, and potential reviewer of many journals and conferences such as IEEE TAP, IEEE AWPL, IET MAP, IET Electronics Letters etc. He is presently handling many research projects in the capacity of Principal Investigator sponsored from different funding agencies like SERB(DST), ISRO, TEQIP etc.

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