High inter-port isolation dual circularly polarized slot antenna with split-ring resonator based novel metasurface

High inter-port isolation dual circularly polarized slot antenna with split-ring resonator based novel metasurface

Int. J. Electron. Commun. (AEÜ) 107 (2019) 146–156 Contents lists available at ScienceDirect International Journal of Electronics and Communications...

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Int. J. Electron. Commun. (AEÜ) 107 (2019) 146–156

Contents lists available at ScienceDirect

International Journal of Electronics and Communications (AEÜ) journal homepage: www.elsevier.com/locate/aeue

Regular paper

High inter-port isolation dual circularly polarized slot antenna with split-ring resonator based novel metasurface Sumantra Chaudhuri, Rakhesh Singh Kshetrimayum ⇑, Ramesh Kumar Sonkar Department of Electronics and Electrical Engineering, Indian Institute of Technology Guwahati, Assam 781 039, India

a r t i c l e

i n f o

Article history: Received 9 January 2019 Accepted 10 May 2019

Keywords: Dual circularly polarized antenna Split ring resonator Slot antenna MIMO Isolation

a b s t r a c t In this paper, a Dual Circularly Polarized (DCP) full-duplex planar slot antenna suitable for C-band applications is presented. The antenna has a single slot with a T-shaped stub that is shared by both the transmitting (T X ) and receiving (RX ) ports which operate at two mutually orthogonal circular polarizations. A metasurface made up of 4 pairs of face-to-face Split Ring Resonators (SRRs) acts as a decoupling network to enhance the isolation between T X and RX ports. The designed antenna has an impedance bandwidth of 1340 MHz which is 28:52% of centre frequency f 0 ¼ 4:71 GHz. The 3 dB axial ratio (AR) bandwidth is 8:97% centred at f AR ¼ 4:44 GHz whereas inter-port isolation is better than 27 dB and reaches up to 36 dB within the usable band. The proposed design also exhibits characteristics suitable for MIMO applications. Ó 2019 Elsevier GmbH. All rights reserved.

1. Introduction In-band-full-duplex (IBFD) communication holds tremendous potential in future wireless communication as it has the capability to improve network throughput. Especially, in military radios, this technique can be used to disrupt the enemy communication while at the same time, continue to receive the intended signals [1]. Furthermore, IBFD can be used to build radios with compact and lighter filters [2]. However, the main challenge in realization of IBFD is the self-interference (SI) of the transmitter, which being orders of magnitude higher than the receiving signal, tends to suppress the latter [2]. A two-port dual polarized (DP) antenna can be used in IBFD communication provided that the inter-port isolation is high enough to prevent the SI from saturating the receiver. However, one major hurdle in mitigating the SI is the mutual coupling (MC) [3] that exists between the co-located transmitting (T X ) and the receiving (RX ) elements of the antenna produced due to surface wave propagation. The literature is abound with different techniques proposed for mitigating SI in DP antennas that can be broadly categorized as dual linearly polarized (DLP) antennas or dual circularly polarized (DCP) antennas. Nawaz and Tekin [4] present a dual linearly polarized microstrip patch antenna where the isolation between T X and RX ports is enhanced by differential feeding of the latter through a hybrid ring coupler. Sun et al. [5] propose an antenna in which the ⇑ Corresponding author. E-mail address: [email protected] (R.S. Kshetrimayum). https://doi.org/10.1016/j.aeue.2019.05.016 1434-8411/Ó 2019 Elsevier GmbH. All rights reserved.

radiator is formed by a square loop with staircase-like sides using two orthogonal feeds. However, both these designs are that of DLP antennas and lack the advantages of a circularly polarized (CP) antenna which are effective resistance to multi-path interference and convenience of orientation of T X and RX antennas. Additionally, CP antennas are effective against Faraday Rotation effect usually seen in the upper layers of the atmosphere [6]. Among the recently reported DCP antennas the one by Chakrabarti [7] comprises of two spatially orthogonal feeds corresponding to two apertures coupled to a patch whose corners have been chamfered for CP operation. The main drawback of [7] is the high-profile structure ( 0:11k0 in height) and narrow impedance bandwidth. The DCP antenna by Shen et al. [8] consists of corner-truncated sequentially fed patch array whose main shortcomings are low interport isolation ( 10 dB), a multi-layer structure and having different resonant frequencies for left-handed circular polarization (LHCP) and right-handed circular polarization (RHCP). Arnaud et al. [9] present a choke horn antenna excited by DCP feed whose main deficiencies are non-planar structure and complicated fabrication process. Since antenna compactness is a major requirement, it becomes imperative to have decoupled networks in DCP antennas to nullify the effects of surface waves. Most of the methodologies feature decoupling networks placed between the radiating elements namely usage of slot on the ground-plane between two patches in proximity [10], placing of a meander-line resonator between patches [11] and presence of a polarization-conversion isolator [12]. The limitations of these approaches are unwanted radiation

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from the slot [10], poor isolation improvement [11] and narrow band of operation [11,12]. This work addresses some of these limitations by designing a strictly duplex antenna using a novel decoupling unit to reduce SI that is easy to fabricate and compact in size. We propose a slot antenna with a T-shaped stub which is shared by both the T X and RX sections. This is similar to the design in [13] with the difference being that in our case the T-section alone of the slot is sufficient in generating CP waves in conjunction with the feed-lines and hence has a simpler design. Additionally, in our approach the isolation between the T X and RX sections is enhanced by means of a SRR based decoupling metasurface between two feed-lines on the opposite surface of the slotted ground plane beneath the FR-4 substrate. This follows the technique similar to the one in [14] where the metasurface suppresses the surface waves in the substrate which is the main cause of higher isolation. The proposed design has the major advantage of being inexpensive and lowprofile and is suitable for radio altimeter applications, being superior to the contemporary ones in terms of bandwidth [15] and polarization diversity [15–17]. The antenna can also be used for 5G applications described in [18] as it has sufficient CP bandwidth at 4:5 GHz.

2. Antenna geometry The proposed antenna has been fabricated on inexpensive FR-4 substrate (er ¼ 4:3; tan d ¼ 0:02) of 1:6 mm thickness with 1 oz copper on both sides. The dimensions of the antenna are 105 mm  50 mm. The exploded view of the antenna is shown in Fig. 1. The antenna has a rectangular slot cut out of the ground plane as depicted in Fig. 2(a). The slot has a T-shaped structure at its base and is excited by two microstrip feed-lines which in turn terminate with T-shaped ends as shown in Fig. 2(b) at the bottom. The width W F1 of the feed is chosen as 3 mm so that the characteristic impedance is 50 X and is hence compatible with standard devices. The feed-lines are separated by a novel metasurface (MS) in the middle that is made up of 4 pairs of SRRs whose slits face each other. The enlarged view of the decoupling MS with 2 pairs of rings is shown in Fig. 3(a). Fig. 3(b) shows the schematic of a single such SRR.

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The dimensions of the slot on the ground are roughly chosen using the equation [19]:



2f

c pffiffiffiffi

er

ð1Þ

where W is the slot dimension, c is the velocity of light in vacuum, f is the operational frequency and er is the effective dielectric constant. From this equation, the initial value of the W s ¼ 23 mm (as in Fig. 2(a)) is obtained which is then optimized using CST Microwave Studio for best performance. The effective width of the slot is further increased by the portion of the T-section AB that yields a high bandwidth as can be seen from the current distribution in Fig. 23. The slot should be long for a leftward shift in resonance frequency as seen in [20]. For this reason Ls is initially chosen as 25 mm but because of poor impedance bandwidth and CP performance at the desired band, it is finally optimized to 43:75 mm which is sufficient to be accommodated within the ground-width Wg. The evolution of the proposed design shown in Fig. 4. The first antenna of Fig. 4(a) has rectangular microstrip feed lines and a square-slotted ground plane. The second design of Fig. 4(b) is very similar to the first one with the only difference being a thin rectangular section on the slotted ground plane. Fig. 4(c) has the additional modification of a T-section on the rectangular slot whereas Fig. 4(d) shows the final prototype which was implemented for design and experimentation. Figs. 5–8 show the jS11 j parameters, ARs, jS12 j parameters and gain of the prototypes versus frequency. It is obvious that Prototype 1 is unsuitable for use as a CP antenna due to very poor impedance bandwidth, high AR P 15 dB and low gain. Prototypes 2 and 3, in spite of having sufficient 3 dB AR bandwidth, suffer from improper impedance matching and poor gain whereas Prototype 4 is found to be most suitable for DCP operation being free of all the drawbacks with the exception of port-to-port isolation which can be remedied by a decoupling unit. Thus it can be concluded that the T-section at the terminal of the feed line is essential for good impedance matching. 3. Design of metasurface As mentioned earlier, the MS is composed of 4 pairs of SRRs. Here, each unit cell can be viewed to be made up of two SRRs

Fig. 1. Exploded view of the antenna.

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Fig. 2. Schematic of the antenna (a) Slotted ground (Lg ¼ 105 mm, W g ¼ 50 mm, LS ¼ 43:75 mm, W S ¼ 34:50 mm, W 1 ¼ 3:125 mm, W 2 ¼ 35:25 mm, T L1 ¼ 13:2 mm, T L2 ¼ 15 mm, W L1 ¼ 1 mm, W L2 ¼ 1 mm) (b) Feed-network (Lg ¼ 105 mm, W g ¼ 50 mm, LF ¼ 41:22 mm, L2 ¼ 12:5 mm, W F1 ¼ 3 mm, W F2 ¼ 1:75 mm).

Fig. 3. Schematic of the metasurface (a) MS (G1 ¼ 2 mm, G2 ¼ G3 ¼ 0:25 mm) (b) SRR (Rr1 ¼ 3:7 mm, Rr2 ¼ 2:2 mm, Rg1 ¼ 0:3 mm, Rg2 ¼ 0:1 mm, Rw1 ¼ 0:7 mm, Rw2 ¼ 0:416 mm).

whose slits face each other as shown in Fig. 9. The resonant frequency of a single split ring resonator is determined using the equation [21]:

1 pffiffiffiffiffiffiffiffiffiffiffi f0 ¼ 2p LT C eq

C eq ¼ ð2Þ

The total inductance LT is given by [22]:

  4l LT ¼ 0:0002l 2:303 log  c d

The equivalent capacitance C eq is given as [21]:

ð3Þ

where l is the length of the SRR, d is the width of the gap between outer ring and inner ring of SRR and c is a constant whose value is 2:451 for a circular SRR.

ðpr 0  gÞ 0 Rw2 h C pul  2 2g

ð4Þ

where r 0 is the average ring-radius, C pul is the capacitance per unit length, Rw2 is the thickness of the inner ring, h is the thickness of substrate and g ¼ 0:5ðRg1 þ Rg2 Þ is the average width of ring splits (as in Fig. 3(b)). The above equations are utilized to arrive at a starting value of ring dimensions which are then further optimized for resonance at desired the frequency. It is to be noted that since each unit cell has two such rings whose currents mitigate each other, the effective

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Fig. 4. Evolution of the proposed antenna (without MS) through four different prototypes.

Fig. 5. jS11 j of the antenna prototypes. Fig. 7. jS12 j of the antenna prototypes.

Fig. 6. Axial ratio of the antenna prototypes.

electrical length is less than the one obtained from the equations Eqs. (2)–(4). Software simulations showed that the parameters most instrumental for optimum performance of this structure are - tilt-angle a of a ring with respect to X-axis (as in Fig. 9), horizontal gap between the two rings within a unit cell (G2 as in Fig. 3(a)), vertical

Fig. 8. Gain of the antenna prototypes.

spacing between two adjacent unit cells (G3 as in Fig. 3(a)) outer ring radius (Rr1 as in Fig. 3(b)) and number of such unit cells. Fig. 10 shows the port-to-port isolation and gain against frequency for different values of a. Here, a is measured in anti-

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Fig. 9. Unit cell of the MS showing two SRRs.

Fig. 10. Variation of (a) jS21 j and (b) gain with respect to frequency at different ring tilt angles a.

clockwise sense for the left SRR and in clockwise sense for the right SRR. Within the AR bandwidth (i.e. 4:245 GHz to 4:635 GHz), the jS21 j values at a ¼ 60 and at a ¼ 0 are very close to each other though the gain at the latter tilt-angle is 0:5 to 1 dBic higher than the same at other angles. Hence a is chosen as 0 which implies that the slits lie along the X-axis. The effects on jS21 j and gain for different values of intra-cell spacing G2 were studied as shown in Fig. 11. This was examined for two close gaps - 0:05 mm and 0:25 mm and a larger spacing

of 4:5 mm. It can be observed from Fig. 11(a) and (b) that a larger spacing between the rings within a unit cell has adverse effects on both inter-port isolation and gain. Therefore, G2 is selected as 0:25 mm as opposed to 0:05 mm for simpler fabrication. The parameter controlling the periodicity of the unit cells is designated as G3 (as in Fig. 3(a)) which is the spacing between two adjacent unit cells along the Y-axis. This value was found to be affecting both the inter-port isolation and AR as shown in Fig. 12(a). It is observed that though a larger value of G3 results

Fig. 11. Variation of (a) jS21 j and (b) gain with respect to frequency for different values of G2 .

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Fig. 12. Variation of (a) jS21 j and (b) AR with respect to frequency for different values of G3 .

from these plots that for the best possible port decoupling, Rr1 should be 3:7 mm and the number of unit cells should be 4. 4. Parametric study

in a minor increase in inter-port isolation, it comes at the expense of a diminished AR bandwidth that has a rightward shift. For this reason G3 was selected as 0:25 mm. The other two parameters influencing the inter-port isolation are the radius of the outer ring Rr1 and the number of unit cells as depicted in Fig. 13 and Fig. 14 respectively. It can be inferred

It was found through several simulations that the most important parameters for optimum antenna performance were: length of the T-section of the feed L2 (represented by GH and KL of Fig. 2(b)), gap between feed and MS G1 (as in Fig. 3(a)), the length of ground Lg and the width of the ground W g (as in Fig. 2). The simulated jS11 j parameters for different values of L2 are shown in Fig. 15. It can be seen that for L2 less than 12:5 mm, the impedance matching degrades within the desired frequency band whereas the resonant frequency shifts right for L2 ¼ 16:0 mm. Hence the most suitable value of L2 is chosen as 12:5 mm. Fig. 16 shows the variation of gain and jS11 j with respect to G1 . It is observed that for the best impedance matching and the most stable gain over entire bandwidth G1 must be equal to 2 mm. Fig. 17 depicts the inter-port isolation and gain of the antenna for different values of Lg . It is obvious from this figure that while there is a minor improvement in the interport isolation the gain improves by 0:5 dBic to 1:2 dBic within the band of 3:9 GHz to 4:6 GHz for Lg ¼ 105 mm. Finally, it is also observed that the W g has an impact on the CP performance of the antenna as shown in Fig. 18. It can be deduced that for W g ¼ 50 mm the best AR bandwidth is obtained as the other values of W g yield reduced usable AR bandwidth.

Fig. 14. jS21 j versus frequency for different numbers of unit cells.

Fig. 15. Simulated jS11 j parameters for different values of L2 .

Fig. 13. jS21 j versus frequency for different values of outer ring radius.

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Fig. 16. Variation of (a) jS11 j and (b) AR with respect to frequency for different values of G1 .

Fig. 17. Variation of (a) jS21 j and (b) AR with respect to frequency for different values of Lg .

Fig. 18. Simulated AR with respect to frequency for different values of W g .

5. Results and discussions The photographs of the fabricated and tested design are shown in Fig. 19. We have chosen to excite only the Port 1 for our radia-

tion pattern and AR measurements and terminated the Port 2 with a matched load during these processes. Because of symmetry and reciprocity, similarly-natured parameters will be obtained if the exciting port is reversed. Fig. 20 shows the S-parameters of the antenna. It is seen from the jS11 j plot that the 10 dB impedance bandwidth is 1340 MHz lying between 4:03 GHz to 5:4 GHz for f 0 ¼ 4:71 GHz. This wideband nature is consistent with that of a planar slot antenna. The same plot also shows the jS21 j of the antenna with and without the MS. The difference in levels of jS21 j in the two cases is the evidence of the filtering action of the MS. The same figure also shows the measured values of jS22 j and jS12 j which are almost same as jS11 j and jS21 j respectively except for minor differences that confirms the symmetric nature of the antenna about its ports. Fig. 21 shows the current distribution on the lowermost pair of SRRs. As evident, because the currents are induced in the mutually opposite direction in the SRRs, their effects are mitigated and hence the surface waves are attenuated while reaching the opposite port. This results in high inter-port isolation. Fig. 22 shows the difference in surface current distribution between the Tsections of the T X and RX ports with and without the MS which further confirms the role of the MS in port decoupling. Fig. 23 shows the surface current distribution on the slotted ground. Since Port 1 has been excited, the surface currents have higher magnitude on the right-hand side of the slot. The vector sum of the current components at each of the phase angles is shown by red colored arrow. Due to the clockwise spinning of

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Fig. 19. Photograph of the fabricated antenna.

band the isolation (with the MS) is better than 27 dB which is a significant improvement of about 10 dB over the same antenna without the MS. The gain of the antenna is fairly constant over its AR bandwidth and ranges from 3 dBic to 3:4 dBic. The xz-plane and yz-plane radiation patterns are shown in Fig. 25. It is seen from Fig. 25(b) that the peak power difference between the left-handed CP (LHCP) and right-handed CP (RHCP) pattern is around 24 dB with the main lobe directed at 28 . This is in accordance with the equation [24]:

XPD ðin dBÞ  10  log

Fig. 20. Plots of simulated and measured S-parameters against frequency.

the tip of the resultant vector LHCP mode is generated at z > 0 plane. Swept frequency AR measurement was done using the technique described in [23]. Fig. 24 shows the 3 dB AR bandwidth and the gain against frequency. The AR bandwidth is 390 MHz which is 8:97% of the centre frequency f AR ¼ 4:44 GHz. At this

ðarÞ2 þ 1 ðarÞ2  1

ð5Þ

where XPD is the cross-polarization discrimination (in dB) and ar is the numeric value of AR. At this particular angle the AR numeric value is 1:008 which is an evidence of good polarization purity. MIMO Performance: Because of good decoupling between the ports it is worthwhile to calculate the envelope correlation coefficient (ECC) as this quantity is an important parameter to check the suitability of the antenna for MIMO performance [25]. The ECC (qe ) is calculated as [25]:

qe ¼

jS11 S12 þ S21 S22 j2 2

ð1  jS11 j  jS21 j2 Þð1  jS22 j2  jS12 j2 Þ

ð6Þ

For the proposed antenna, this parameter was calculated to be less than 0:18 which satisfies the requirement of being less than 0:5 for practical purposes. Also the diversity gain given by [26]:

Fig. 21. Surface current distribution on the lowermost SRR pairs of the MS at f AR ¼ 4:44 GHz.

Fig. 22. Surface current distribution on the feed-lines without the MS and with the MS at f AR ¼ 4:44 GHz.

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Fig. 23. Surface current distribution on the slotted-ground at f AR ¼ 4:44 GHz for different phase-angles.

Fig. 24. Axial ratio and gain against frequency.

qffiffiffiffiffiffiffiffiffiffiffiffiffiffi DG ¼ 10 1  q2e

ð7Þ

is found to be almost stable at 10 dB. Thus it can be concluded that the antenna performs satisfactorily as a MIMO antenna. The plot of qe and DG (in dB) is shown in Fig. 26. The Table 2 gives the comparison of performances between the proposed antenna and other contemporary antennas. It is seen that this design performs better in almost all the parameters excepting gain. The moderate gain is due to low Front-to-Back Ratio (FBR) which is a characteristic of slot antenna [27] and is compensated by fairly omni-directional radiation pattern. The gain, however, is comparable to other recently reported antennas suggested for radio altimetry applications [28,29] and is better than the MIMO antennas presented in [30,31]. The simulated efficiency of the antenna is greater than 68% and reaches up to 79% within the 10 dB impedance band. Gain improvement. It is well-known that lossy substrates with high tan d can yield low antenna gains [32,33]. Hence it becomes necessary to examine the antenna performance for different substrates and explore the possibilities of augmenting the gain. Fig. 27 shows the simulated efficiency of the antenna when different substrates (tabulated in Table 1) are used. The substrates chosen have close relative permittivities for a meaningful comparison. As expected, the substrate FR-4 with highest loss tangent (tan d) is the least efficient when used in the antenna compared to the low-

Fig. 25. Radiation pattern at f ¼ 4:44 GHz.

loss ones. Fig. 28 depicts the effect on antenna gain due to usage of different substrates. Here again, it is confirmed that a low-loss substrate yields higher gain. As mentioned earlier, the gain is moderate due to the bidirectional pattern of the slot antenna. In such a case usage of a metallic reflector on one side is suggested to improve the gain [34,35]. However reflectors have been known to adversely affect the CP performance as seen in [36]. Therefore the functionality of the antenna in terms of both gain and AR bandwidth (with Taconic RF-43 substrate) is examined by placing a copper reflector (135 mm  65 mm) at a distance of hp ¼ 7:5 mm beneath the

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Fig. 26. Plot of ECC (qe ) and DG versus frequency.

Fig. 28. Gain of the antenna versus frequency for different substrates. Fig. 27. Efficiency of the antenna versus frequency for different substrates.

Table 1 Comparison of properties of different substrates. Substrate

Relative permittivity (er )

FR-4 Arlon AD-450 Rogers TMM4 Preperm PPE440 Taconic RF-43

Loss tangent (tan d)

Thickness

4:3 4:5 4:5 4:4

0:02 0:0035 0:002 0:009

1:6 mm 1:52 mm 1:52 mm 1:6 mm

4:3

0:0033

1:57 mm Fig. 29. Schematic of the antenna showing the copper reflector in yz-plane.

Table 2 Comparison of proposed antenna with other recently reported antennas. Ref.

Pol.

Inter-port isolation

10 dB B.W.

AR B.W.

Antenna size

Gain

Pol. diversity

[7] [8] [9] [16] [37] [38] This work

Circular Circular Circular Circular Linear Linear Circular

15 dB to 28 dB 12 dB to 25 dB 10 dB to 40 dB – 27 dB to 29 dB 29 dB to 37 dB 27 dB to 36 dB

13:23% and 9:52% 23:69% and 18:5% 4:87% 109:3% and 28:5% 10:5% 17:1% 28:52%

7:8% 12:5% and 14:7% 4:87% 3% to 37:3% – – 8:97%

0:73 k0  0:73 k0  0:0036 k0 3:54 k0  1:69 k0  0:2768 k0 2:18 k0  2:18 k0  0:22 k0 1:0 k0  0:76 k0  0:022 k0 1:0 k0  0:50 k0  0:03 k0 0:84 k0  0:66 k0  0:0124 k0 1:58 k0  0:75 k0  0:024 k0

3:5 dBic to 4:3 dBic 4 dBic to 10:7 dBic 5 dBic to 6:4 dBic 2:57 dBic to 5:78 dBic 2:7 dBi to 5:6 dBi 4:7 dBi to 5:2 dBi 2:9 dBic to 3:6 dBic

Yes Yes Yes No Yes Yes Yes

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Fig. 30. AR bandwidth of the antenna for two different types of substrates.

microstrip feed lines as shown in Fig. 29. With this arrangement, the gain is substantially increased to > 7:5 dBic as seen from Fig. 28. The CP performance of the antenna remains largely unaffected and the AR bandwidth undergoes a minor increase as seen in Fig. 30. In practical situations requiring high gain, the antenna may be placed at a distance hp away from a metallic wall which would act as a reflector. 6. Conclusions We have presented a DCP slot antenna comprising of a novel metasurface based decoupling network for isolation enhancement between the ports. The measured results closely follow the simulated parameters. The proposed antenna, with its high interport isolation and low AR within its 10 dB impedance bandwidth, is appropriate for C-band applications such as radio altimeters. Also because of its low ECC, it is suitable as a diversity antenna for 5G MIMO applications. Declaration of Competing Interest The authors declared that there is no conflict of interest. Appendix A. Supplementary material Supplementary data associated with this article can be found, in the online version, at https://doi.org/10.1016/j.aeue.2019.05.016. References [1] Riihonen T, Korpi D, Rantula O, Valkama M. On the prospects of full-duplex military radios. In: International conference on military communications and information systems (ICMCIS). p. 1–6. [2] Hong S, Brand J, Choi JI, Jain M, Mehlman J, Katti S, et al. Applications of selfinterference cancellation in 5G and beyond. IEEE Commun Mag 2014;52 (2):114–21. [3] Venkatakrishnan SB, Alwan EA, Volakis JL. Wideband RF self-interference cancellation circuit for phased array simultaneous transmit and receive systems. IEEE Access 2018;6:3425–32. [4] Nawaz H, Tekin I. Compact dual-polarised microstrip patch antenna with high interport isolation for 2.5 GHz in-band full-duplex wireless applications. IET Microwaves Antennas Propag 2017;11(7):976–81. [5] Sun K, Yang D, Chen Y, Liu S. A broadband commonly fed dual-polarized antenna. IEEE Antennas Wirel Propag Lett 2018;15(5):747–50. [6] Gao SS, Luo Q, Zhu F. Circularly polarized antennas. 1st ed. John Wiley & Sons; 2013.

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