Stepless solid-state controls for battery-powered dc electric vehicles

Stepless solid-state controls for battery-powered dc electric vehicles

W. MCMurray STEPLESS SOLID-STATE CONTROLS FOR BATTERY-POWERED DC ELECTRIC VEHICLES W. McMurray, Electrical Engineer, General Electric Company, Schene...

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W. MCMurray

STEPLESS SOLID-STATE CONTROLS FOR BATTERY-POWERED DC ELECTRIC VEHICLES W. McMurray, Electrical Engineer, General Electric Company, Schenectady, New York, USA SUMMARY The operating principle of stepless dc controllers of the chopper type is described, with application to control series dc motors in either the motoring or braking mode. Such systems are suitable for driving battery powered vehicles. Methods of adjusting the time ratio of the chopper to regulate its output are discussed. While transistor choppers are suitable for low-power drives, thyristor choppers are employed for higher power. Auxiliary circuit branches required for commutation in a thyristor chopper are analyzed by their function. Many of the different circuit arrangements that have been devised are reviewed and compared. ZUSAMMENFASSUNG Die Arbeitsweise einer Steuerung fur stufenlose Gleichstrompulswandler fur Hauptschlussmotoren in Fahr- und Bremsrichtung wird berictet. Solche Steuerungen sind fur Batteriegespeiste Triebfahrzeuge geeignet. Methoden werden berictet uber die Einstellung des Einschaltverhaltnisses zur Steuerung des Laststromes. Transistoren sind fur kleinere und Thyristoren fur grossere Leistungen geeignet. Die Funktion der Kommutierungstromkreise wird untersucht. Verschiedene, eingesetzte Stromkreise werden besprochen und verglichen.

PRINCIPLE OF STEP LESS DC CONTROL Stepless control of power to a dc load can be obtained by a chopper circuit using solid-state switching devices such as power transistors or thyristors. The controller is a means of matching a voltage source (or sink) Ed to a current sink (or source) Id, as sketched in figure 1 where an ideal swjtch S is shown in place of the semiconductor device and its accessories. If necessary, a capacitor CF or an inductance LF may be added to maintain the voltage or current, respectively, nearly constant under transient conditions. The direction of power flow depends on whether the current Id is returned to the positive terminal P (dashed line) or the negative terminal N (chain line) of the voltage Ed.

r---~--...,~ -- - - - - - - - --, Figure 1: Ideal dc chopper circuit

L

1.:

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VOLTAGE SOURCE/SINK

CURRENT

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Operation of the ideal chopper is explained by the waveforms in figure 2. When the switch S is open (time TOFF)' the current Id must flow through the diode 0 and the voltage Ed appears across the switch. When the switch S is closed (time TON)' the current naturally commutates to the path through S and diode 0 blocks the voltage Ed. Thus, as the switch opens and closes repetitively with a period T, the current Id is chopped into alternate pulses through S or 0, as indicated in figure 2(A). The potential of node Z in figure 1 switches from the positive terminal P of the voltage source to the negative terminal N, as seen in figure 2(B). The average potential of node Z is indicated by the dashed lines in figure 2(B)-(F).

(A)

~

0

I sI

0

Iso

S

0

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(8)

(C)

(D)

(E)

(F)

Figure 2: (A) (B) (C) (D) (E) (F)

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--t------±---t--±-~--±-I---I-±--f-~-. Waveforms in ideal dc chopper circuit Division of current between diode 0 and switch S Voltage at node Z: solid line, instantaneous value; dashed line, average value Voltage change with constant frequency Voltage change with constant "on" time Voltage change with constant "off" time Voltage change with variable frequency, "on" time and "off" time

W. McMurray

When the current flows from the positive terminal P, the current sink absorbs power at an average voltage Ed TON/T. Current pulses of duration TON flow from the voltage source through switch S, yielding an average current Id TON/T and an average power EdId TON/T. Similarly, when the current flows from the negative terminal N, an average power EdId TOFF/T is delivered from the current source to the voltage sink. Thus, in either case, the power is determined by the ratio of the time TON or TOFF to the total period T. For this reason, the tec~~ique is often called "time ratio control". A chopper behaves as an adjustable-ratio dc autotransformer; the potential of node Z can be varied between zero and Ed by controlling the time ratio. Four methods by which the pulse timing depicted in figure 2(B) can be altered to obtain a new average potential at Z are shown in figure 2(C)-(F). In figure 2(C), the ratio TON/TOFF has been changed but the total period T remains the same. Since in most practical circuits TON or TOFF cannot be reduced below a certain minimum, a restriction of the time ratio range is implied when the total period remains constant. Therefore, for battery-powered vehicles variable-frequency control methods are generally employed. Some types of choppers operate with a near constant value of TON. In this case, the frequency must be increased as shown in figure 2(D) ~o produce the same time ratio change. If it were the off time TOFF that must be constant, then the frequency would be reduced as shown in figure 2(E). For a more flexible circuit, both TON and TOFF can be adjusted to obtain the desired time ratio at some suitable frequency, as illustrated in figure 2(F). CHOPPER-CONTROLLED DC MOTOR DRIVES The chopper principle can be applied as a stepless controller for a dc machine in either motoring or generating (braking) operation/l,2/. Connections suitable for a battery-powered vehicle driven by a series-field dc motor are sketched in figure 3. The advantages of series motors for electric traction are well known. The motoring connection is shown in figure 3(A). Comparing this with figure 1, it is seen that the battery is in the position of the vOltage source and the motor is in the position of the current sink. The inductance of the field and armature acts as a filter choke. The machine generates a counter emf of polarity indicated in figure 3(A) and does not naturally behave as a current sink, but nearconstant current operation can be obtained by suitable feedback control. Several methods of braking may be employed, in addition to mechanical brakes which are essential for safety and at low speeds where electrical methods "fade out". For regenera423

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Figure 3: Application of chopper to drive dc series motor

(A)

(A) Motoring

1 1

op: SJ

(B)

(B) Regenerative braking

(C)

R

(C) Dynamic braking

(D)

(D) Plug braking

1 1

tive braking, figure 3(B), the motor armature is reversed with respect to the field and relocated in the current source position. The reconnection is generally effected by contactors. The battery acts as a voltage sink and accepts the kinetic energy regenerated by the motor in braking. At high speeds, a diode indicated by dashed lines in figure 3(B) may be necessary to limit the armature voltage. Regenerative braking is preferred for road vehicles. Besides conserving battery energy, it avoids the problem of dissipating heat when descending long grades. Use of a chopper makes regenerative braking feasible in practice. Dynamic braking is shown in figure 3(C). Energy regenerated by the motor is dissipated in resistor R. The motor and controller operate in a manner similar to regenerative braking, as can be seen if a capacitor is visualized across the braking resistor and diode 0 is retained. The capacitor acts transiently like a voltage sink. Although dynamic braking is less desirable, it can be used without a chopper controller and where the source cannot accept energy. 424

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Sometimes, the method known as plug braking is employed. In figure 3(0), the machine remains in the current sink position of figure 1, but the field is reversed so that the armature voltage reverses and circulates current through the diode Dp. The kinetic energy is dissipated as heat in the armature windings. During plugging, the chopper must operate with a very low duty cycle (TON/T small) to limit the field current. Because of motor heating, plugging is not suitable when extended braking periods are required. CONTROL OF CHOPPER FREQUENCY AND RIPPLE CURRENT When motoring, figure 3(A), assume that the machine generates a counter emf Ea and neglect the resistance of the windings. Then, the voltage (Ed- Ea) appears across the inductance LF of the armature and field when the switch S is conducting and the voltage (-Ea) appears across LF when diode 0 is conducting. This alternating voltage causes the motor current to deviate by ± 1 6 from the average value Id' as indicated in figure 4 for the steady-state condition where Ea = Ed TON/T. It is easily shown that (1 )

The repetition frequency f should be high enough so that the inductance of the motor itself provides sufficient filtering of the current. However, the frequency should be no higher than necessary, since the switching losses are proportional to frequency. If a tolerable value of the ripple current l a is selected, (1) can be rearranged to yield the frequency as a function of the voltage ratio Ea/Ed' equal to TON/T f

1 T

(2 )

The frequency function (2) is a parabola, as sketched in figure 5, and is maximum when Ea/Ed = 1/2. (3 )

19ure 4: Voltag e and current in dc chopper with counter emf load

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VOLTAGE RATIO EolEd TIME RATIO TOM IT

Figure 5: Frequency variation for constant ripple current amplitude

It is possible to design an oscillator that adjusts its frequency in this manner according to a voltage demand signal. However, this type of control does not restrict the current and must be overridden by a current limit regulator. A better method is to use the signal set by the operator as a current reference which is compared with the actual motor current. The comparator operates to turn on the switch when the current falls below (Id - 1 0 ) and turn it off when the current reaches (Id + 1 <5 )/3,4/. This bangbang control d1rectly produces the waveforms in figure 4 and adjusts the frequency and time ratio to satisfy (2). A current limit is naturally obtained by rest.ricting the demand signal.

POWER TRANSISTOR CHOPPER CIRCUIT Low-power drives may use transistors as the chopper switch, avoiding commutating capacitors and other auxiliary components that are required by thyristors. However, a capacitor is generally needed to limit the rise of voltage as the transistor turns off, reducing its dissipation and preventing second breakdown. As indicated in figure 6, a diode DA limits discharge of the capacitor when the transistor is turned on again, so that the stored energy O.5C Ed 2 is dissipated in resistor R instead of the transistor/5/. In practice, the inductance Lo of the battery and the cables is in the supply loop. This is useful in that it limits the rise of current as the device switches on.

Figure 6: DC chopper using a power transistor switch S

T R

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However, inductance delays the transfer of current from the source loop including capacitor C to the coast loop through diode D after the transistor has turned off. In the process, the capacitor is charged above the battery voltage by Id 1Lolc and the energy 0.5 LoId 2 originally stored in Lo is lost in resistor R as the capacitor discharges to the level Ed again. The total power PR in resistor R may be substantial at high power and high frequency: (4)

REVIEW OF THYRISTOR CHOPPER CIRCUITS When the switch is a thyristor, auxiliary means are required to turn it off. Essentially, a pre-charged capacitor is connected to apply reverse voltage across the thyristor for at least the turn-off time. Other components are required to control the charge on the capacitor, reverse its polarity, and time the start of commutation. The simplest method has an LC circuit across the thyristor, as shown in figure 7/6/. The waveforms illustrate certain problems common to thyristor choppers. (A) Circuit diagram

(B) Flux/ampereturn diagram for inductance L

(C) VOltages

(D) Currents

Figure 7:

Basic chopper using thyristor SCR 427

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List of Problems 1. 2. 3. 4.

5. 6.

The "on" time is fixed, so that the only method of control is to vary th e frequency. The capacitor reversal current is superimposed over the load current in the thyristor, increasing its duty. The rise di /dt in the thyristor and the fall di/dt in the diode are very high, causing high switching losses. The capacitor vOltage oscillates during the "off" in terval, so that the commutating energy depends upon when the thyristor is next fired. When the loa d is light, the recover y interval becomes longer and limits the repetition rate. There is no way to charge the capacitor other than through th e load , which may pose a problem in some cases, as in figure 3(C) .

Many different circuits have been devised to overcome thes e problems, depending on the needs of the application. This section will review thyristor chopper operation from a functional viewpoint. An attempt will be made to classify various commutating arrangements and evaluate them as con trollers for electric ve hicles. Chopper Circuit with Separate Functional Branches A circuit where all of the listed problems are overcome by the addition of auxiliary thyristors is shown in figure 8. However, such complexity is not practical and it is desirable to omit or replace most of the auxiliary thyristors with diodes if possible. At least one auxiliary thyristor must be retained to control the "on" time. The figure shows a separate inductance in series with each component. This serves as a reminder that parasi tic inductance is important during the fast switching transient s , and also shows where lumped inductors may be located. As will be discussed, it is desirable to minimize the inductance in certain paths, and to limit the number of lumped inductors.

Figure 8: Chopper circuit with thyristors in separate functiona l branches

Ed

The usual sequence of action is shown in figure 9, where the loops active during each interval are marked. Descrip tive names are given to each loop and the corresponding intervals. The behavior depends on the total inductance in 428

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Figure 9 : Sequence of active l oops in operation of chopper , figure 8

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(G)

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W. McMurray

each loop and the firing time of the thyristor that closes the loop. The various functions will now be discussed. Boost (precharge of capacitor). - The loop through SCR-O in fIgUre 9(A) overcomes problem 6, but is needed only during starting or special load conditions. Often, the capacitor can be charged through the load by firing SCR-A. Such action is automatic if SCR-A is replaced by a diode. Power. - The power loop is the path from the source through the load, filter and main SCR, and is active when SCR is "on", as shown in figure 9(B) and (C). Reversal (of capacitor voltage). - The capacitor charge must be reversed to obtain the proper polarity for turning off the main thyristor. A separate path through SCR-B avoids problem 2, and inductance in the loop is required to avoid an impulse of infinite current for infinitesimal time. Reversal during the power interval is most usual, figure 9(B), but with isolation provided by thyristors in the other loops, it can occur any time between the end of the overcharge interval and the start of the extinction interval. Figure 9(F) shows reversal during the coast interval, allowing the power interval to be as short as desired. In circuits where reversal is restricted to the power interval, the minimum "on" time is the reversal time. Extinction (of current in SCR). - Thyristor SCR-A provides the tim~ng function to overcome problem 1 and is fired to apply the reversed capacitor voltage to turn off the main thyristor, as indicated in figure 9(C). In circuits where SCR-A is replaced by a diode, the timing function is assumed by SCR-B and extinction immediately follows reversal. It is desirable to have some inductance in the loop to limit the fall di/dt and recovery losses of the thyristor. Recharge (of capacitor). - The. recharge interval, figure 9(0), immediately follows extinction. In figure 7, it is designated the "recovery" interval. The load current recharges the capacitor through thyristor SCR-A. With maximum current, the capacitor must be sized to maintain reverse voltage on thyristor SCR for its turn-off time. Rebound (fast recharge of capacitor). - With light load current, the recharge interval becomes long. Since it is not desirab~e to refire the main thyristor before the capacitor is fully charged, this restricts the repetition rate and constant frequency operation is not feasible. This is listed as problem 5, and is solved by providing a rebound loop through thyristor SCR-C, see figure 9(0). However, it is usual to avoid such complexity by replacing SCR-C with a diode and increasing the capacitance to allow for charge diverted through the diode. There must be inductance in the loop to limit the amount diverted.

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If a rebound diode is connected directly across thyristor SCR or SCR-B, the reverse-bias time for a given capacitance can increase. However, the reverse voltage is limited to the diode drop and is followed by very high dv/dt, requiring a snubber such as shown in figure 6. Hence, the direct inverse-parallel connection is less desirable. For most battery-powered vehicles, constant frequency operation is not employed, very light load conditions are not encountered, and a rebound path can be omitted. Coast.- The coast loop is the path through the load, filter, and coasting diode D, as marked on figure 9(E), (F) and (G). Overcharge (of capacitor).- Transfer of current from the recharge path to the coast path is not instantaneous because of inductance in the overcharge loop shown in figure 9(E). This is the same phenomenon that was discussed with a transistor chopper. The capacitor is charged above the source voltage by an amount proportional to the current Id. (5)

With low voltage and high current, overcharge may be desirable. However, it is usual to minimize the overcharge. Thus, no inductors should be in series with diode D, SCR-A or capacitor C. The source can be stiffened by a filter capacitor and wiring inductance should be minimal. Despite such precautions, some overcharge will occur in heavycurrent circuits and a blocking device should be in the loop to hold the overcharge on the capacitor, preventing the oscillations listed as problem 4. In figure 8, thyristor SCR-A serves this purpose. Turn-on (of main thyristor).- Again, transfer of current from the coast path to the power path is rate-limited by inductance in the turn-on loop shown in figure 9(G). Inductance here is desirable to limit the rise di!dt in SCR, which is the fall di/dt in diode D. This reduces the switching losses of both devices, listed as problem 3. Practical Chopper Circuits Derived from Figure 8 The circuit of figure 10 is obtained from figure 8 if we neglect any boost requirement, retain inductors only where they are desirable (plus the inevitable source inductance) and replace auxiliary thyristors with diodes, except SCR-B. Three alternative locations for the optional rebound diode D-C are indicated by dashed lines. Replacement of thyristors SCR-A and SCR-C by diodes restricts the timing of loop closures and results in overlap between the reversal, extinction, rebound and recharge If SCR-B is fired at the same time as SCR to intervals. obtain a short "on" time, then the reversal interval also overlaps the turn-on interval. This may increase the 431

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Figure 10: Practical chopper derived from figure 8.

capacitance required, but the trade-off is justified in view of the high cost of thyristors with their firing circuits. The arrangement of figure 11 allows full reversal of the capacitor charge before the extinction interval commences. However, the inductance is included in the overcharge loop, which is to be avoided unless overcharge is desired.

Figure 11: Circuit allowing full capacitor reversal.

Circuits with Main Thyristor in Reversal Loop If problem 2 is ignored, the other problems can be solved by a circuit in which the reversal current passes through the main thyrlstor as well as an auxiliary device. Early in the development of choppers, the linear reactor of figure 7 was replaced by a saturating reactor/6/. A similar effect and better timing capability can be achieved by a pair of thyristors and a linear reactor, figure 12(A). Since only one thyristor is really necessary for timing, the other can be replaced by a diode, figure 12(B) or (C). For all circuits of this family, reversal is restricted to some part of the power interval. In figure 12(C), reversal occurs when the main thyristor is fired, aggravating its di/dt problem. Thus, figure 12(B) is preferred, where SCRB is fired just before SCR is to be turned off, and extinction automatically follows reversal. In all of these arrangements, rebound diode D-C can be added directly across SCR, as indicated, or in series with an inductance. To minimize inductance in the overcharge loop, the scheme of figure 13(A) is an improvement over figure 12(C). The 432

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to-c

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L-____-+________________

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. SCR-B L ._ . _ . _ . (B)

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Figure 12: Circuits with main thyristor in reversal loop (A) Two auxiliary thyristors (B) Extinction automatically follows reversal (C) Reversal automatically follows turn-on auxiliary thyristor and diode in figure 13(A) cannot be interchanged as in figure 12(B) since a diode D-A would prevent reversal of the capacitor voltage. A new problem arises with the circuits of figures 12(C) and 13(A), because the capacitor overcharge is converted into an undercharge by swing-back through diodes D-B and D, see the loop indicated on figure 13(A) . The (+) sign in (6) becomes (-). Since a situation where the commutating ability decreases as the load current increases is intolerable, the swing-back must be blocked by using a thyristor SCR-B as in figure 13 (B) / 7/ , or by inserting an extra diode D-S, as in figur e I3(C). In all the v ariations of the figure 8 circuit family, a swing-back problem does not arise. Circuits with Multi-Purpose Inductance All of the ~esired functions of commutating inductance in a chopper can be obtained by locating a single reactor L in the reversal, extinction, rebound and turn-on loops, 5ut not the overcharge loop, as shown in figure 14. This circuit uses no more components than the circuit of figure 10, and has the advantage that both thyristors SCR and SCR-B have the same turn-off tim e . As in other circuits, source inductance has the effect of 433

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(A) Undercharge swing-back loop

(A)

c

(E) Two auxiliary thyristors

(8)

c

(C) Extra blocking diode

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c

Figure 13: Circuits illustrating "swing-back to undercharge" problem and its solution

o Figure 14: Chopper with mUlti-purpose inductance

-- -

--, I

O-A

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W. McMurray

overcharging the capacitor. When SCR-B is fired to close the reversal loop through inductance Ll' part of the overcharge is lost by temporary conduction of diode D; there is a "swing-back loop" through C, SCR-B, SCR, D, Lo and the source. This effect can be avoided by adding a small inductance L2 in series with thyristor SCR-B such that (6)

CONCLUSIONS Battery-powered vehicles propelled by a dc motor can be controlled in a stepless manner by chopper controllers. These are much more efficient than resistor controllers during acceleration and also allow the use of regenerative braking. The behavior of the chopper during braking is similar to the motoring mode, and suitable switching enables the same equipment to be used for both functions. While low-power drives may use transistors, thyristors are preferred for high power. Many different circuits have been devised for thyristor choppers. Typically, two thyristors, two or three diodes, a commutating capacitor and one or more inductances are employed. The major art in design is selecting the most suitable configuration for the application, particularly the location of the inductances. It is desirable to minimize the inductance in certain loops and to carefully choose the inductance in other loops. An attractive arrangement, figure 14, has the major inductance in a path that is common to both the main and auxiliary thyristors. REFERENCES /1/ J. Gouthiere, J. Gregoire, and H. Hologne: Thyristor Choppers in Electric Traction, ACEC Review (1970), No. 2, pp. 45- 6 7 . /2/ P . Knapp: Solid State Regulating Units for Motoring and Braking DC Traction Vehicles, Brown Boveri Rev. 57 (1970), No. 6-7, pp. 252-270. /3/ J.V. Byrne and J.G. Lacy: Compatible Controller-Motor System for Battery-Electric Vehicle, Proc. lEE 117 (1970), pp. 369 - 376 . /4/ K. Heumann: Pulse Control of DC and AC Motors by Silicon-Controlled Rectifiers, IEEE Trans. Commun. Electron, 83 (1964), pp. 390-399. /5/ H.F. Weber: Pulse-Width Modulation DC Motor Control, IEEE Trans. Ind. Electron. Contr. Instrum. IECI-12 (1965), pp. 24-28. /6/ W. McMurray: SCR DC to DC Power Converters, IEEE Trans. Commun. Electron. 83 (1964), pp. 198-203. /7/ M. Meyer: Thyristoren in der technischen Anwendung. Band 1: Stromrichter mit erzwungener Kommutierung, Siemens Co., Germany (1967), p. 81.

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